RF power amplifier

ABSTRACT

In one embodiment, an RF power amplifier includes a first transistor and a second transistor in parallel, wherein a gate of the first transistor and a gate of the second transistor are configured to be driven by an RF source. A third transistor comprising a drain is operably coupled to both a source of the first transistor and a source of the second transistor. A control circuit is operably coupled to a gate of the third transistor and configured to alter a gate-to-source voltage of the third transistor, thereby altering a drain current of each of the first transistor and the second transistor, thereby altering an output power of the RF power amplifier.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is a continuation in part of U.S. patentapplication Ser. No. 16/255,269, filed Jan. 23, 2019, which is acontinuation in part of U.S. patent application Ser. No. 16/211,961,filed Dec. 6, 2018, which is a continuation in part of U.S. patentapplication Ser. No. 15/787,374, filed Oct. 18, 2017, which is acontinuation in part of U.S. patent application Ser. No. 15/667,951,filed Aug. 3, 2017 (now U.S. Pat. No. 10,217,608), which is acontinuation of U.S. patent application Ser. No. 15/384,904, filed Dec.20, 2016 (now U.S. Pat. No. 9,729,122), which is a continuation in partof U.S. patent application Ser. No. 15/046,585, filed Feb. 18, 2016 (nowU.S. Pat. No. 9,525,412), which is a continuation in part of U.S. patentapplication Ser. No. 14/734,053, filed Jun. 9, 2015 (now U.S. Pat. No.9,306,533), which claims the benefit of U.S. Provisional PatentApplication No. 62/118,552, filed Feb. 20, 2015.

The present application further claims the benefit of U.S. ProvisionalPatent Application No. 62/670,990, filed May 14, 2018. U.S. patentapplication Ser. No. 16/255,269 further claims the benefit of U.S.Provisional Patent Application No. 62/620,781, filed Jan. 23, 2018. U.S.patent application Ser. No. 16/211,961 further claims the benefit ofU.S. Provisional Patent Application No. 62/595,222, filed Dec. 6, 2017.U.S. patent application Ser. No. 15/787,374 further claims the benefitof U.S. Provisional Patent Application No. 62/409,635, filed Oct. 18,2016. U.S. patent application Ser. No. 15/046,585 further claims thebenefit of U.S. Provisional Patent Application Ser. No. 62/117,728,filed on Feb. 18, 2015. The foregoing references are incorporated hereinby reference in their entireties.

BACKGROUND

Semiconductor wafer fabrication can use plasma processing to manufacturesemiconductor devices, such as microprocessors, memory chips, and otherintegrated circuits and devices. Plasma processing involves energizing agas mixture by introducing RF energy. This gas mixture is typicallycontained in a vacuum chamber, also called plasma chamber, and the RFenergy is introduced through electrodes or other means in the chamber.In a typical plasma process, the RF generator generates power at an RFfrequency and this power is transmitted through RF cables to the plasmachamber. To provide efficient transfer of power from the RF generator tothe plasma chamber, a matching network is used.

In the plasma process and other applications, there is a need to varythe output power of the RF generator. A previously known method for aClass E amplifier uses the variable DC bus to control and vary the RFoutput power. This technique is inherently slow, however, since the DCbus cannot respond to fast control loop demands driven by the output RFpower needs. They also use pulse width control in the RF stage, whichmakes the requirement difficult at higher frequencies. The maindisadvantage of previous methods is the use of a very expensive fast DCbus power supply, which is used to power the Class E or Class D RFamplifiers. The power amplifiers, by themselves, can be efficient, butif one further considers the power losses of the fast bus, they are moreexpensive and less efficient overall. As a result, there is need for aless expensive and more efficient method for varying the output power ofan RF generator.

BRIEF SUMMARY

In one aspect, the present disclosure is direct to an RF power amplifiercomprising a first transistor and a second transistor in parallel,wherein a gate of the first transistor and a gate of the secondtransistor are configured to be driven by an RF source; a thirdtransistor comprising a drain operably coupled to both a source of thefirst transistor and a source of the second transistor; and a controlcircuit operably coupled to a gate of the third transistor andconfigured to alter a gate-to-source voltage of the third transistor,thereby altering a drain current of each of the first transistor and thesecond transistor, thereby altering an output power of the RF poweramplifier.

In another aspect, the present disclosure is direct to a method ofamplifying RF power, the method comprising coupling a first transistorand a second transistor in parallel; driving, by an RF source, a gate ofthe first transistor and a gate of the second transistor; coupling adrain of a third transistor to both a source of the first transistor anda source of the second transistor; coupling a control circuit to a gateof the third transistor; and altering, by the control circuit, agate-to-source voltage of the third transistor, thereby altering a draincurrent of each of the first transistor and the second transistor,thereby altering an output power of the RF power amplifier.

In another aspect, the present disclosure is direct to a method ofmanufacturing a semiconductor, the method comprising placing a substratein a plasma chamber configured to deposit a material layer onto thesubstrate or etch a material layer from the substrate; energizing plasmawithin the plasma chamber by coupling RF power from an RF source intothe plasma chamber to perform a deposition or etching, the RF sourcecomprising an RF power amplifier comprising a first transistor and asecond transistor in parallel, wherein a gate of the first transistorand a gate of the second transistor are configured to be driven by an RFsource; a third transistor comprising a drain operably coupled to both asource of the first transistor and a source of the second transistor;and a control circuit operably coupled to a gate of the thirdtransistor; and altering a gate-to-source voltage of the thirdtransistor, thereby altering a drain current of each of the firsttransistor and the second transistor, thereby altering an output powerof the RF power amplifier.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing summary, as well as the following detailed description ofthe exemplary embodiments, will be better understood when read inconjunction with the appended drawings. It should be understood,however, that the invention is not limited to the precise arrangementsand instrumentalities shown in the following figures:

FIG. 1A is a schematic representation of a first embodiment of an RFimpedance matching network.

FIG. 1B is a representation of a first embodiment of a virtual ground.

FIG. 2 is a schematic representation of a second embodiment of an RFimpedance matching network.

FIG. 3 is a schematic representation of a third embodiment of an RFimpedance matching network.

FIG. 4 is a schematic representation of a fourth embodiment of an RFimpedance matching network.

FIG. 5 is a schematic representation of a fifth embodiment of an RFimpedance matching network.

FIG. 6 is a schematic representation of a sixth embodiment of an RFimpedance matching network.

FIG. 7 is a schematic representation of a seventh embodiment of an RFimpedance matching network.

FIG. 8 is a schematic representation of an eighth embodiment of an RFimpedance matching network.

FIG. 9 is a schematic representation of a ninth embodiment of an RFimpedance matching network.

FIG. 10 is a schematic representation of a tenth embodiment of an RFimpedance matching network.

FIG. 11 is a schematic representation of a first switching circuitaccording to one embodiment.

FIG. 12 is a schematic representation of a second switching circuitaccording to another embodiment.

FIG. 13 illustrates parasitic capacitances on a switching circuitaccording to one embodiment.

FIG. 14 is a graph of a switched waveform according to one embodiment.

FIG. 15 is a block diagram of a switching circuit according to oneembodiment.

FIG. 16 is a schematic representation of a third switching circuitaccording to one embodiment.

FIG. 17 is a schematic representation of a fourth switching circuit.

FIG. 18 is a switching circuit having a driver circuit for switching aPIN diode according to one embodiment

FIG. 19 is a timing diagram for the driver circuit of FIG. 18.

FIG. 20 is a schematic of a system having parasitic capacitancecompensation circuits for discrete capacitors of an EVC according to oneembodiment.

FIG. 21 is a graph showing output capacitance variation for a typicalSiC high-voltage MOSFET according to one embodiment.

FIG. 22 is a graph showing a power curve for a typical SiC high-voltageMOSFET according to one embodiment.

FIG. 23 is a graph of a maximum power dissipation of a typical SiChigh-voltage MOSFET according to one embodiment.

FIG. 24 is a simplified schematic of a system utilizing anelectronically variable capacitor according to one embodiment.

FIG. 25 is a schematic of a high-power RF switch for a discretecapacitor of the electronically variable capacitor of FIG. 24.

FIG. 26 is a schematic representing the high-power RF switch of FIG. 25in the ON state.

FIG. 27 is a schematic representing the high-power RF switch of FIG. 25in the ON state during the positive cycle.

FIG. 28 is a schematic representing the high-power RF switch of FIG. 25in the ON state during the negative cycle.

FIG. 29 is a schematic representing the high-power RF switch of FIG. 25in the OFF state.

FIG. 30 is an equivalent circuit for the schematic of FIG. 29 with theaddition of a tuning circuit according to one embodiment.

FIG. 31 is a schematic providing a practical circuit layout for the forhigh-power RF switch of FIG. 25 according to one embodiment.

FIG. 32 is a second embodiment of the high-power RF switch utilizingtransistors instead of diodes according to one embodiment.

FIG. 33 is a third embodiment of the high-power RF switch for increasedhigh voltage according to one embodiment.

FIG. 34 is a simulation circuit for a high power-RF switch when in theOFF state according to one embodiment.

FIG. 35 is a simulation circuit for a high power-RF switch when in theON state according to one embodiment.

FIG. 36 is a waveform of a signature of the RF current flowing throughone of the discrete capacitors of the EVC according to one embodiment.

FIG. 37 is a block diagram of a topology for a universal RF amplifieraccording to one embodiment.

FIG. 38 is a schematic of an RF amplifier for a Class E amplifieraccording to one embodiment.

FIG. 39 is a schematic of an RF amplifier for a Class AB amplifieraccording to one embodiment.

FIG. 40 is a block diagram of a topology for a quadrature RF amplifierfor Class E operation according to one embodiment.

FIG. 41 provides a graph providing a detailed view of the timing of thegate drives according to one embodiment.

DETAILED DESCRIPTION

The description of illustrative embodiments according to principles ofthe present invention is intended to be read in connection with theaccompanying drawings, which are to be considered part of the entirewritten description. In the description of embodiments of the inventiondisclosed herein, where circuits are shown and described, one of skillin the art will recognize that for the sake of clarity, not alldesirable or useful peripheral circuits and/or components are shown inthe figures or described in the description. Moreover, the features andbenefits of the invention are illustrated by reference to the disclosedembodiments. Accordingly, the invention expressly should not be limitedto such disclosed embodiments illustrating some possible non-limitingcombinations of features that may exist alone or in other combinationsof features; the scope of the invention being defined by the claimsappended hereto.

As used throughout, ranges are used as shorthand for describing each andevery value that is within the range. Any value within the range can beselected as the terminus of the range. In addition, all references citedherein are hereby incorporated by reference in their entireties. In theevent of a conflict in a definition in the present disclosure and thatof a cited reference, the present disclosure controls.

Matching Network

As discussed above, to provide efficient transfer of power from the RFgenerator to the plasma chamber, a matching network is used. The purposeof the matching network is to transform the plasma impedance (usually alow value) to a value suitable for the RF generator. In many cases,particularly in wafer fabrication processes, the RF power is transmittedthrough 50 Ohm coaxial cables and the system impedance of the RFgenerators is also 50 Ohm. On the other hand, the impedance of theplasma, driven by the RF power, varies and this impedance must betransformed to non-reactive 50 Ohm (i.e., 50+j0) for maximum powertransmission. RF matching networks perform this task of continuouslytransforming the plasma impedance to 50 Ohm for the RF generator. An RFmatching network can include one or more variable capacitors and amicroprocessor-based control circuit to control the capacitance valuesof the variable capacitors. The value and size of the variablecapacitors are influenced by the power handling capability, thefrequency of operation, and the impedance range of the plasma chamber.

The predominant variable capacitor in use in RF matching networks is theVacuum Variable Capacitor (VVC). Electronically Variable Capacitor (EVC)technology, however, is emerging as an alternative, as EVCs can beswitched more quickly. Faster switching enables faster matching, whichenables faster manufacturing. EVCs comprise discrete capacitors that areswitched in or out to alter the total EVC capacitance.

In the embodiments of an RF impedance matching network disclosed herein,the matching network is configured as a “PI” type matching network. Bythis configuration, the switching of the variable capacitance componentsand variable inductance components (sometimes referred to collectivelyas variable components or reactive components) occurs in the shunt ofthe matching circuit. Thus, unlike in other matching networks, such as“L” type matching networks, the exemplified matching network switchesits reactive components to a ground and not in series. There are novariable inductors or capacitors in the series leg.

In these embodiments, an RF impedance matching network includes an RFinput configured to couple to an RF source, the RF source having a fixedRF source impedance; an RF output configured to couple to a load, theload having a variable load impedance; a transformation circuit coupledto the RF input and configured to provide a transformed impedance thatis less than the fixed source impedance; and a PI circuit having a firstshunt circuit in parallel to the RF input and a second shunt circuit inparallel to the RF input and the RF output. The first shunt circuitincludes a first shunt variable component providing a first variablecapacitance or inductance; and a first virtual ground coupled to thefirst shunt variable component and a ground. The second shunt circuitincludes a second shunt variable component providing a second variablecapacitance or inductance; and a second virtual ground coupled to thesecond shunt variable component and the ground.

In general, the circuit configurations are tailored to either aninductive or capacitive load impedance of the chamber. Whether oneconfigures a shunt circuit as an inductor or a capacitor will depend onfactors such as frequency of operation, power, and the appropriatephysical size of the element. For example, smaller inductors will beeasier to package and layout, with lower heat dissipation that is easierto control. Specific embodiments will be described in more detail below.

Turning in detail to the drawings, FIG. 1A illustrates an RF impedancematching network 100 according to a first embodiment of the invention.The matching network 100 includes an RF input 160 configured to coupleto an RF source 110, and an RF output 170 coupled to a load 120. Asensor 162 is coupled at the RF input 160 between the RF source 110 andthe matching network 100. In the exemplified embodiment, the load 120 isa plasma chamber for semiconductor device fabrication. The semiconductordevice can be microprocessor, a memory chip, or another type ofintegrated circuit or device. In other embodiments, the load 120 can beany load of variable impedance that can utilize an RF matching network.

In the exemplified embodiment, the RF impedance matching network 100serves to help maximize the amount of RF power transferred from the RFsource 110 to the plasma chamber 120 by matching the impedance at the RFinput 160 to the fixed impedance of the RF source 110. The matchingnetwork 100 can consist of a single module within a single housingdesigned for electrical connection to the RF source 110 and plasmachamber 120. In other embodiments, the components of the matchingnetwork 100 can be located in different housings, some components can beoutside of the housing, and/or some components can share a housing witha component outside the matching network 100.

As is known in the art, the plasma within a plasma chamber 120 typicallyundergoes certain fluctuations outside of operational control so thatthe impedance presented by the plasma chamber 120 is a variableimpedance. Since the variable impedance of the plasma chamber 120 cannotbe fully controlled, an impedance matching network may be used to createan impedance match between the plasma chamber 120 and the RF source 110.

Moreover, the impedance of the RF source 110 may be fixed at a set valueby the design of the particular RF source 110. Although the fixedimpedance of an RF source 110 may undergo minor fluctuations during use,due to, for example, temperature or other environmental variations, theimpedance of the RF source 110 is still considered a fixed impedance forpurposes of impedance matching because the fluctuations do notsignificantly vary the fixed impedance from the originally set impedancevalue. Other types of RF sources may be designed so that the impedanceof the RF source may be set at the time of, or during, use. Theimpedance of such types of RF sources is still considered fixed becauseit may be controlled by a user (or at least controlled by a programmablecontroller) and the set value of the impedance may be known at any timeduring operation, thus making the set value effectively a fixedimpedance.

The RF source 110 may be an RF generator of a type that is well-known inthe art to generate an RF signal at an appropriate frequency and powerfor the process performed within the plasma chamber 120. The RF source110 may be electrically connected to the RF input 160 of the RFimpedance matching network 100 using a coaxial cable or similar means,which for impedance matching purposes would have the same fixedimpedance as the RF source 110.

The plasma chamber 120 can include a first electrode and a secondelectrode, and in processes that are well known in the art, the firstand second electrodes, in conjunction with appropriate control systems(not shown) and the plasma in the plasma chamber 120, enable one or bothof deposition of materials onto a substrate and etching of materialsfrom the substrate.

The sensor 162 is configured to monitor the RF signal output from the RFsource 110. The sensor 162 can monitor an RF input parameter or multipleRF input parameters at the RF input 160. The sensor 162 can be anysensor configured to detect a parameter at the RF input 160. The inputparameter can be any parameter measurable at the RF input 160 sufficientfor operating the matching network 100. In the exemplified embodiment,the sensor 162 detects the voltage, current, and phase at the RF input160 of the matching network 100. The sensor 162 can provide analoginformation to a control unit 180, where it is converted to a digitalform. Based on the RF input parameter detected by the RF input sensor162, the control unit 180 can determine the variable load impedance. Thecontrol unit 180 can further calculate the necessary switching positionson the shunt legs such that the proper inductance and capacitance isprovided by the variable components 144, 154 for creating an impedancematch. That is, the control unit 180 can determine a first shuntcapacitance value for the first shunt variable capacitance component 144and a second shunt capacitance value for the second shunt variablecapacitance component 154 to create an impedance match at the RF input160. The control unit 180 can then send a control the driver circuit 185to alter a first shunt variable capacitance of the first shunt variablecapacitance component 144; and alter a second shunt variable capacitanceof the second shunt variable capacitance component 154 based on a secondcontrol signal received from the control unit 180. The match need not bea perfect matching of impedance. For example, an impedance match canresult in 10% or less RF power reflected back to the RF source.

Most of the inductive and capacitive components used in the discussedembodiments can be designed on a ceramic substrate or some othermaterial such as Rogers material that can withstand the temperatureexcursions. Particularly at higher power levels and large currents acapacitive array or/and inductive array may be specifically packaged tosatisfy the thermal issues associated with current passing through theshunt elements and series elements at higher power levels. Many of thesecomponents will be either air cooled or water cooled, depending on theparticular architecture used for a specific load.

Transformation Circuit

The matching network 100 of the exemplified embodiment includes both aPI circuit 135 and a transformation circuit 130. The fixed impedancetransformation circuit 130 is located at the front-end of the matchingnetwork 100. The transformation circuit 130 is coupled to the RF input160 and enables the impedance at cross-section A-A looking back towardRF input 160 to be lowered to a value that is less than the real part ofthe fixed RF source impedance, thus providing a desired loweredimpedance at the input of the PI circuit 135 (the PI circuit input 132)that is less than the fixed impedance of the RF source 110. In thisembodiment, the RF source impedance is fixed at 50 Ohms, and RF power istransmitted through coaxial cables which also have a fixed impedance of50 Ohms. In other embodiments, the fixed impedance can be of a differentvalue. In preferred embodiments, the lowered impedance is less than thereal part of the load impedance (R_(L)).

The transformation circuit 130 includes an inductor L1 (sometimesreferred to as a transformation inductor) and a capacitor C1 (sometimesreferred to as a transformation capacitor). In this embodiment, thetransformation inductor L1 is in series with the RF input 160, and thetransformation capacitor C1 is in parallel to the RF input 160 andcoupled to a chassis ground GND. In other embodiments, the configurationis reversed, with the capacitor in series with the RF input 160, and theinductor in parallel to the RF input 160 and coupled to the chassisground GND. The configuration chosen depends on the remaining PI circuit135 and the configuration that prevents the DC component of the load 120returning to the RF source 110. The transformation circuit 130 isconnected to a chassis ground GND (not the virtual grounds, which willbe discussed below). In other embodiments, the chassis ground GND can beanother type of ground.

In the exemplified embodiment, the transformation circuit 130 isconfigured to lower the impedance from 50 Ohms at the RF input 160 toapproximately 15 Ohms at the PI circuit input 132 (the input of the PIcircuit 135), with no imaginary part (or minimal imaginary part). Thus,for example, the output impedance (Z_(o)) of the transformation circuit130 is approximately 15 Ohms+j0. The PI circuit 135 of the matchingnetwork 100 is designed for this reduced input impedance, referred to asZ_(in) in FIGS. 1A-10. The interface between the output of thetransformation circuit 130 and the PI circuit 135 is designated by lineA-A.

The transformation circuit 130 can lower the voltage stresses on thematching network 100. Thus, high voltage stress on switches S11 to S1Nand S21 to S2N will be lowered. Such lowered stress enhances thefunctioning of switches such as RF FET switches, PIN diodes, andinsulated-gate bipolar transistors (IGBTs).

This lowered stress can be better understood by the following examples.In the first example, there is no transformation circuit. A 5,000 WattsRF generator (RF source) has a 50 Ohms output impedance (R_(source)) anda frequency of 27.12 MHz that is provided at the RF input 160 of thematching network 100. The matching network 100 is perfectly tuned to 50Ohms at its input and has no reactive components present. The voltage(V=√{square root over (PR)}) therefore will be √{square root over ((5000W)(50 Ohms))}) or 500 V rms. The current (1=V/R) will be 500 V rms/50Ohms, or 10 A rms.

In the second example, a transformation circuit lowers the impedanceprovided at the input 132 of the PI circuit 135 to 15 Ohms. The voltage(V=√{square root over (PR)}) will now be √{square root over ((5000 W)(15Ohms))}) or 273.9 V rms, and the current (1=V/R) will be 273.9 V rms/15Ohms, or 18.3 A rms. Thus, the current increases by a factor of 1.83,while the voltage decreases by a factor of 1.83. The active componentsof the PI circuit 135 more easily handle current than voltage. Thus, thetransformation circuit's reduction of voltage causes less stress on theactive components. A designer can choose for his convenience appropriateimpedance reduction factor. That reduction factor depends on chamberload impedance and its real part (Z_(L)=R_(L)+/−jX_(L)). In preferredembodiments, the reduced input impedance Z_(in) is less than the realpart of the load impedance (R_(L)).

PI Circuit

The PI circuit 135 of FIG. 1A includes a series capacitor C3 in serieswith the RF input 160 and coupled between the transformation inductor L1and the RF output 170. The series capacitor C3 can decouple the DCcomponent to help prevent the DC component of the load 120 fromreturning to the RF source 110. The PI circuit 135 further includes aseries inductor L2 in series with the RF input 160 and coupled betweenthe series capacitor C3 and the RF output 170. The PI circuit 135further includes a first shunt circuit 140 in parallel to the RF input160 and a second shunt circuit 150 parallel to the RF input 160.

The first shunt circuit 140 includes a first shunt capacitor C_(DC) anda first shunt padding inductor L3 _(P). The first shunt capacitor C_(DC)can decouple the DC component of the plasma coming back toward theswitches S11, S12, S1N, such that the switches are not burdened withlarge components of the DC voltage. The first shunt circuit 140 furtherincludes a first shunt variable inductive component 144 comprising (a) aplurality of first shunt inductors L31, L32, L32 coupled in parallel tothe first shunt padding inductor L3 _(P), and (b) a plurality of firstshunt switches S11, S12, S1N coupled to the plurality of first shuntinductors L31, L32, L32 and configured to connect and disconnect each ofthe plurality of first shunt inductors L31, L32, L32 to a first virtualground 142. Also included is a first shunt ground capacitor C1 _(gnd)coupled between the second virtual ground and the ground GND. The firstshunt ground capacitor C1 _(gnd) and a first shunt ground inductanceinherent to the first virtual ground 142 can resonate in series toprovide the lowest possible impedance from the virtual ground to theground GND. The first shunt circuit 140 further includes a first shuntgalvanic return resistor R1 _(G) coupled between the first virtualground 142 and the ground GND. The first shunt ground capacitor C1_(gnd) and the first shunt galvanic return resistor R1 _(G) are coupledin parallel.

The second shunt circuit 150 includes a second shunt variablecapacitance component 154 comprising (a) a plurality of second shuntcapacitors C21, C22, C2N coupled in parallel, and (b) a plurality ofsecond shunt switches S21, S22, S2N coupled to the plurality of secondshunt capacitors C21, C22, C2N and configured to connect and disconnecteach of the plurality of second shunt capacitors C21, C22, C2N to asecond virtual ground 152. The circuit further includes a paddingcapacitor C2 _(P) coupled in parallel to at least one of the pluralityof second shunt capacitors C21, C22, C2N, the padding capacitor C2 _(P)coupled to the second virtual ground 152. The circuit further includes asecond shunt ground capacitor C2 _(gnd) coupled between the secondvirtual ground 152 and the ground GND, and a second shunt galvanicreturn resistor R2 _(G) coupled between the second virtual ground 152and the ground GND. The second shunt ground capacitor C2 _(gnd) and thefirst shunt galvanic return resistor R2 _(G) are coupled in parallel.The capacitors, inductors, and resistors discussed herein can refer toany components of suitable design to provide capacitance, inductance,and resistance, respectively. In preferred embodiments, the matchingnetwork is designed such that, for a particular load, there is minimalpossible variation of capacitors and inductors, thereby allowing thefewest possible switches. Padding capacitors and padding inductors arecapacitors and inductors that do not need to be switched.

The variable inductance components discussed herein can refer to aplurality of shunt inductors and coupled switches, and is sometimesreferred to as an inductive array or simply a variable inductor.Similarly, the variable capacitance components discussed herein canrefer to a plurality of shunt capacitors and coupled switches, and issometimes referred to as a capacitive array or simply as a variablecapacitor. A variable inductance component can refer to any structurecapable of providing a variable inductance, and a variable capacitancecomponent can refer to any structure capable of providing a variablecapacitance. For example, the variable capacitance component can be anelectronic variable capacitor (EVC), as described in U.S. Pat. No.7,251,121. By these components, the capacitances and inductancesprovided in the shunt legs can be controlled such that the combinedimpedances of the RF impedance matching network 100 and the plasmachamber match, or at least substantially match, the fixed impedance ofthe RF source 110. A first shunt variable inductance component 144 and asecond shunt variable capacitance component 154 are shown in FIG. 1A.

Virtual Ground

As discussed above, the switches are not connected directly to chassisground GND but to a virtual ground 142, 152. FIG. 1B shows an embodimentof the first virtual ground 142. The same or a similar structure can beused for virtual ground 152 and the other virtual grounds disclosedherein. In the exemplified embodiment of FIG. 1B, the virtual ground 142is an aluminum plate with embedded tubes 147. The virtual ground 142 iswater cooled, with water (H₂O) flowing in and out of the embedded tubes147. The virtual ground can include thermally conductive isolation padsor paste between the virtual ground surface 145 and the chassis GND toconduct heat and to separate the virtual ground from the chassis groundGND. The pad 143 is not electrically conductive. In other embodiments,the virtual ground 142, 152 can be any electrically conductive componentthat is physically separated from a ground. For example, the virtualground can be arranged vertically or include fins. Further, the virtualground can include other cooling means. For example, at low powerapplications, the virtual ground can be air cooled. Standard heatsinking methods can be applied.

In the exemplified embodiment of FIG. 1B, nine switches S11-S19 aremounted on the surface 145 of the virtual ground 142. The nine switchesS11-S19 correspond with switches S11, S12, and S1N of FIG. 1A, since the“N” can refer to any total number of switches (or any total number ofinductors or capacitors as “N” is used elsewhere in the drawings). Inother embodiments, more or less switches can be used, depending on therequired accuracy of the variable inductance (or variable capacitance inother embodiments).

Switchable and padding components L31-L39 and L3 _(P) (corresponding toL31, L32, L3N, L3 _(P) of FIG. 1A) can also be mounted on the surface145 of the first virtual ground 142, as shown in FIG. 1B. Theexemplified embodiment uses nine switchable inductors, though, as withthe switches discussed above, any number of switchable inductors (orswitchable capacitors depending on the embodiment) can be used. Further,the other virtual grounds disclosed herein can be configured similarly.Thus, the switchable and padding components C21, C22, C2N, C2 _(P) ofthe second shunt circuit 150 can be mounted on a surface of the secondvirtual ground 152 in a manner similar to the surface 145 of virtualground 142 shown in FIG. 1B. Further, similar virtual grounds can beused for the matching network embodiments shown in FIGS. 2-10. Forexample, virtual ground 242 of FIG. 2 can use a virtual groundconfiguration similar to that shown in FIG. 1B, but where capacitorsC11, C12, C1N, C1 _(P) are mounted on the surface of the virtual ground242 instead of inductors L31-L39 and L3 _(P).

As shown in FIG. 1A, the virtual ground 142, 152 can be connected to acommon RF chassis ground GND via a ground capacitor C1 _(gnd), C2_(gnd). The virtual ground 142, 152 has an inherent inductance (L_(gnd))that is generally small. This inherent inductance can connect in serieswith the ground capacitor to create a series resonant circuit. Theseries resonant circuit acts as a short at the frequency of operation.There is also a galvanic return resistor designated as R1 _(G) or R2_(G) that has a much larger value than the series resonance impedance ofL_(gnd) and C_(gnd). Thus, the galvanic connection does not pass themain RF current.

By using a virtual ground for each shunt circuit of variable components,each branch return of RF-switched current can go to one point ground.Since the shunt branch RF current can be very large, they will be mucheasier to control and to stream them away from, for example, FET gatedriving low voltage circuitry. Further, noise immunity and RFinterference within the matching network will be much easier to control.

By the virtual ground, the switches in a shunt circuit can be connectedto a platform from which heat can be better controlled and extracted ifneeded. The RF currents flowing from the tuning capacitors or inductors(e.g., L31, S21) will always flow into the virtual ground 142. Thevirtual ground 142 can also reduce the coupling capacitance from theback side of the switches and their respective FETs to a ground. Eachvariable capacitive or inductive component 144, 154 can have a separatevirtual ground 142, 152 to further reduce the capacitive cross talkamong the switches. The separation of switched currents in the virtualground can help reduce the jitter noise in the switches as well as crosstalk among the switches. These currents can be very difficult to controlat very high RF power levels. The virtual ground can ease the design ofhigh power switches.

After full layout of the circuit, one can determine the inductance ofthe ground connections. A capacitance can be determined for some verysmall impedance from virtual ground to the chassis ground GND bycalculating the needed capacitance C_(gnd). In a preferred embodiment,the capacitor C_(gnd) has a very low ESR (equivalent series resistance)and should withstand several kilovolts of voltage breakdown in case ofovervoltage occurrence on the RF switches. Choosing the propercomponents can prevent the network from being under high voltage stress.The embodiment shown in FIG. 1A is useful for an inductive load chamberand powers below 5 kW.

In the matching network 100 of FIG. 1A and certain other embodimentsdiscussed hereafter, a bypass capacitor C_(DC) (or C4) forms part of theshunt circuit 140 for the purpose of decoupling the DC voltage that maybe reflected from the chamber load 120. This capacitance is not part ofthe matching network 100 in that this capacitance is not used formatching purposes. The capacitor C_(DC) can sustain high voltage swingscoming back from the load 120 and can pass very large RF currents. Forinstance, at RF power P=5 kW, these currents may be in the order of 100A rms. The bypass capacitor C_(DC) can be in series with the variableshunt inductance and can prevent the DC voltage returning from thechamber 120 from reaching the RF switches.

In the exemplified embodiment, the switches use field effect transistors(FETs). In other embodiments, the switches use PIN/NIP diodes, a MicroElectro Mechanical (MEM) switch, a solid state relay, bipolartransistors, insulated-gate bipolar transistors (IGBTs), and the like.In the exemplified embodiment, each switch turns a capacitor or inductorON or OFF to control the overall capacitance or inductance provided bythe variable components 144, 154, thus enabling the matching network 100to provide variable capacitance and variable inductance in the shuntlegs. In alternative embodiments, a switch can be configured to controlmore than one capacitor or inductor.

The matching network 100 includes one or more RF switch driver circuits185. The driver circuits 185 are configured to switch the plurality ofswitches S11, S12, S1N, S21, S22, S2N. Each of the driver circuits 185can include several discrete driving circuits, with each discretedriving circuit configured to switch one of the switches.

The matching network 100 further includes a control unit 180. Thecontrol unit 180 is the brains of the RF impedance matching network 100as it receives multiple inputs from sources such as the sensor 162 andthe driver circuits 185 and makes calculations necessary to determinechanges to the variable capacitance and inductance components 144, 154,and delivers commands to these components 144, 154 to create theimpedance match. The control unit 180 can be of the type that iscommonly used in semiconductor fabrication processes, and thereforeknown to those of skill in the art.

The control unit 180 can be configured with an appropriate processorand/or signal generating circuitry to provide an input signal forcontrolling the driver circuits 185. The control unit 180 of thematching network 100 can be a standard DSP- and FPGA-based architecture.The control unit 180 can house several other circuits, including anovervoltage conditioning circuit 182 for switching off all the activeswitches in the case of overvoltage at the output of the match. Theovervoltage circuit 182 can indicate to the control board when to gointo the shutdown condition.

In the exemplified embodiment, the control unit 180 includes aprocessor. The processor may be any type of properly programmedprocessing device, such as a computer or microprocessor, configured forexecuting computer program instructions (e.g. code). The processor maybe embodied in computer and/or server hardware of any suitable type(e.g. desktop, laptop, notebook, tablets, cellular phones, etc.) and mayinclude all the usual ancillary components necessary to form afunctional data processing device including without limitation a bus,software and data storage such as volatile and non-volatile memory,input/output devices, graphical user interfaces (GUIs), removable datastorage, and wired and/or wireless communication interface devicesincluding Wi-Fi, Bluetooth, LAN, etc. The processor of the exemplifiedembodiment is configured with specific algorithms to enable matchingnetwork 100 to perform the functions described herein.

A power supply (not shown) can be connected to the driver circuits 185,control unit 180, and sensor 162 to provide operational power, at thedesigned currents and voltages, to each of these components.

The inductive and capacitive shunt designs in the PI configurationenable low voltage stresses on the variable components. High voltagestress is particularly hard on active FET switches that must switchlarge potentials as well as large currents at the power levels on theorder of 5 kW. Since the disclosed embodiments do not switch any seriescomponents in these PI configurations, they are fixed in this matchingnetwork 100, and therefore there are lower voltages on the shuntcapacitive or inductive components. This will be shown later in atabular form.

At lower frequencies the inductors may be discrete since they will haveinherently larger values. At higher frequencies such as 13.56 MHz, 27.12MHz, 40.68 MHz, and 60 MHz, the inductors can be made by a method calledspiral inductors and printed on a ceramic substrate.

FIG. 2 is a schematic representation of a second embodiment of an RFimpedance matching network 200. As will be described, this embodimentuses variable capacitance components 244, 254 in both shunt legs. As inFIG. 1A, the matching network 200 includes an RF input 260 configured tocouple to an RF source 210 and an RF output 170 configured to couple toa load 220, a transformation circuit 230, and a PI circuit 235.

The transformation circuit 230 is again coupled to the RF input 260 andconfigured to provide a transformed impedance that is less than thefixed source impedance. The transformation capacitor C1, however, is inseries with the RF input 260, and the transformation inductor L1 is inparallel to the RF input and coupled to the chassis ground GND.

The first shunt circuit 240 is in parallel to the RF input 260. Thecircuit 240 includes a first shunt variable capacitance component 244comprising (a) a plurality of first shunt capacitors C11, C12, C1Ncoupled in parallel, and (b) a plurality of first shunt switches S11,S1, S13 coupled to the plurality of first shunt capacitors C11, C12, C1Nand configured to connect and disconnect each of the plurality of firstshunt capacitors C11, C12, C1N to a first virtual ground 242. The firstshunt circuit 240 further includes a padding capacitor C1 _(P) (“firstshunt padding capacitor”) coupled in parallel to at least one of theplurality of first shunt capacitors C11, C12, C1N, the first shuntpadding capacitor C1 _(P) coupled to the first virtual ground 242; acapacitor C1 _(gnd) (“first shunt ground capacitor”) coupled between thefirst virtual ground 242 and the ground GND; and a resistor R1 _(G)(“first shunt galvanic return resistor”) coupled between the firstvirtual ground 242 and the ground GND.

The second shunt circuit 250 is also in parallel to the RF input 260.Similar to the first shunt circuit 240, the second shunt circuit 250includes a second shunt variable capacitance component 254 comprising(a) a plurality of second shunt capacitors C21, C22, C2N coupled inparallel, and (b) a plurality of second shunt switches S21, S22, S2Ncoupled to the plurality of second shunt capacitors C21, C22, C2N andconfigured to connect and disconnect each of the plurality of secondshunt capacitors C21, C22, C2N to a second virtual ground 252. Thesecond shunt circuit 250 further includes a padding capacitor C2 _(P)(“second shunt padding capacitor”) coupled in parallel to at least oneof the plurality of second shunt capacitors C21, C22, C2N, the secondshunt padding capacitor C2 _(P) coupled to the second virtual ground252; a capacitor C2 _(gnd) (“second shunt ground capacitor”) coupledbetween the second virtual ground 252 and the ground GND; and a resistorR2 _(G) (“second shunt galvanic return resistor”) coupled between thesecond virtual ground 252 and the ground GND.

The matching network 200 further includes a series inductor L2 in serieswith the RF input 260 and coupled between the transformation capacitorC1 and the RF output 270. The embodiment of the matching network 200shown in FIG. 2 is useful for inductive and capacitive loads. Since ithas only one inductor in the series leg, it is efficient. It is usefulfor high power applications that are less than 10 kW.

FIG. 3 is a schematic representation of a third embodiment of an RFimpedance matching network 300. In this embodiment, variable inductors344, 354 are used in both shunt legs. As in the previous figures, thematching network 300 includes an RF input 360 configured to couple to anRF source 310 and an RF output 370 configured to couple to a load 320, atransformation circuit 330, and a PI circuit 335.

Similar to the embodiment shown in FIG. 1A, the transformation inductorL1 is in series with the RF input 360, and the transformation capacitorC1 is in parallel to the RF input 360 and coupled to a chassis groundGND. Also similar to FIG. 1A, the first shunt circuit 340 includes afirst shunt padding inductor L3 _(P); a plurality of first shuntinductors L31, L32, L3N coupled in parallel to the first shunt paddinginductor L3 _(P); a plurality of first shunt switches S11, S12, S1Ncoupled to the plurality of first shunt inductors L31, L32, L3N andconfigured to connect and disconnect each of the plurality of firstshunt inductors L31, L32, L3N to a first virtual ground 342; a firstshunt ground capacitor C1 _(gnd) coupled between the second virtualground 352 and the ground GND; and a first shunt galvanic returnresistor R1 _(G) coupled between the first virtual ground 342 and theground GND. The second shunt circuit 350 is configured similar to thefirst shunt circuit 340. The matching network 300 further includes afirst series capacitor and a second series capacitor coupled in seriesbetween the RF input 360 and the RF output 370.

The embodiment of the matching network 300 shown in FIG. 3 has only oneinductor in the series leg and therefore is more efficient. It is usefulfor high power applications up to more than 5 kW. It is useful forinductive loads.

FIG. 4 is a schematic representation of a fourth embodiment of an RFimpedance matching network 400. This embodiment is similar to theembodiment shown in FIG. 3, however, the first series capacitor C2 iscoupled between the RF input 460 and the first shunt circuit 440, thesecond series capacitor C3 is coupled between the first shunt circuit440 and the second shunt circuit 450, and a single capacitor C4 iscoupled in series with the padding inductor L4 _(P) in the second shuntcircuit 450. The embodiment of the matching network 400 shown in FIG. 4has only one inductor in the series leg and therefore is useful inhigher power matching designs, up to approximately 10 kW. The seriescapacitor C3 is moved away from high current in the load loop, whichimproves efficiency and therefore usefulness for high RF power matching.

FIG. 5 is a schematic representation of a fifth embodiment of an RFimpedance matching network 500. This embodiment is similar to theembodiment shown in FIG. 1A, however, the series inductor L2 is replacedwith a series capacitor C3, a series inductor L4 is added, and singlecapacitor C_(DC) in the first shunt circuit 140 of FIG. 1A iseliminated. The embodiment of the matching network 500 shown in FIG. 5is useful at lower powers up to 5 kW.

FIG. 6 is a schematic representation of a sixth embodiment of an RFimpedance matching network 600. This embodiment is similar to theembodiment shown in FIG. 1A, however, the inductor L1 and capacitor C1of the transformation circuit 630 are reversed, the series capacitor C3of FIG. 1A is eliminated, and the positioning of the first and secondshunt circuit 650 s is reversed, such that the first shunt circuit 640has the plurality of switching capacitors C21, C22, C2N, and the secondshunt circuit 650 has the plurality of switching inductors L41, L42,L4N. The embodiment of the matching network 600 shown in FIG. 6 isuseful for medium-to-high powers and inductive loads. Since it has onlyone inductor in the series leg, it can provide good efficiency.

FIG. 7 is a schematic representation of a seventh embodiment of an RFimpedance matching network 700. This embodiment is similar to theembodiment shown in FIG. 2, however, a series inductor L3 is coupledbetween the transformation capacitor C1 and the first shunt circuit 740.The embodiment of the matching network 700 shown in FIG. 7 has twoinductors in the series leg, and therefore is used for lower powersbelow 5 kW.

FIG. 8 is a schematic representation of an eighth embodiment of an RFimpedance matching network 800. This embodiment is similar to theembodiment shown in FIG. 6, however, the inductor L1 and capacitor C1 ofthe transformation circuit 830 are reversed, a series inductor L2 isadded between the transformation inductor L1 and the first shunt circuit840, and the series inductor L2 between the first and second shuntcircuits 840, 850 is replaced with a series capacitor C3. The embodimentof the matching network 800 shown in FIG. 8 is useful for capacitivechamber loads. There are two inductors in the series leg and it isuseful for applications below 5 kW.

FIG. 9 is a schematic representation of a ninth embodiment of an RFimpedance matching network 900. This embodiment is similar to theembodiment shown in FIG. 2, however, an additional series inductor L3 iscoupled between the second shunt circuit 950 and the RF output 970. Theembodiment of the matching network 900 shown in FIG. 9 is useful forcapacitive chamber loads. There are two inductors in the series leg andtherefore more losses. It is useful for applications below 5 kW.

FIG. 10 is a schematic representation of a tenth embodiment of an RFimpedance matching network 1000. This embodiment is similar to theembodiment shown in FIG. 8, however, the series inductor L3 is coupledbetween the first shunt circuit 1040 and the second shunt circuit 1050,and the series capacitor C3 is coupled between the second shunt circuit1050 and the RF output 1070. The embodiment of the matching network 1000shown in FIG. 10 is useful for inductive chamber loads and formedium-to-low RF power applications. It is noted that, where componentsin one embodiment are similarly situated to components in anotherembodiment, the function and characteristics of those components can besimilar, though a detailed discussion of those components is notrepeated for that particular embodiment.

In Table 1, simulated results for typical matching conditions are shownfor the matching network 100 shown in FIG. 1A. The impedance loadconditions are assumed as typical chamber impedances. Values for seriescoupling capacitor C3 (2 nF) and series inductance L2 (8 uH) werespecifically chosen to satisfy the circuit equations under the matchingconditions. R_(L) is representative of the real part of the loadimpedance. P_(out) is the delivered RF power to the load resistor R_(L).The example components were chosen for delivering 5 kW of RF power to aplasma chamber. The simulations show that the matching network cansatisfy all the load impedance conditions for a typical process.

TABLE 1 Matching Network of FIG. 1A Simulation Results. L31, C21, L32,C22, L3N L2 C2N I_(C3) I_(L2) I_(L) V_(L) R_(L) X_(L) C3 (var.) (fixed)(var.) A, A, A, kV, P_(out) Ω Ω nF uH uH pF rms rms rms rms kW 1 31 22.5 8 3,080 9.8 15.3 66.4 1.6 5 2 38 2 3.1 8 2,640 11.3 13.3 45 1.4 5 539 2 5 8 2,400 9.9 8.0 30.5 0.8 5.8 10 39 2 5 8 2,300 13.3 6.8 20 0.75.1

Software was used to determine the circuit components values for thevariable shunt elements L3, C2 using the assigned load impedance values.The software then calculated the currents and voltages. As is shown, thematching network can be designed with reasonable values for thecapacitors and inductors. The simulation was carried out at thefrequency f=2 MHz, and therefore the components values reflect thatfrequency. The assumed power was 5 kW. One could choose other value forthe variable shunt elements L3, C2 to attempt to have even smallercurrents and voltages in the variable shunt elements L3, C2.

Switching Circuits

FIGS. 11-17 below discuss switching circuits that can be utilized in theabove matching networks. For example, each of switches S11 to S1N andS21 to S2N in FIG. 2 can utilize one of the switching circuits discussedbelow to switch capacitors C11 to C1N and C21 to C2N, respectively, thusenabling electronically variable capacitance. Similarly, the switchingcircuits can switch the inductors L31 to L3N of FIG. 1 to enablevariable inductance. Accordingly, switches S11 to S19 of FIG. 1B canutilize one of the switching circuits discussed below.

The switching circuits discussed below can also be used outside of RFmatching networks, and outside the realm of high frequency switching.For example, certain switching circuits may be used as a form of highvoltage relay at a low frequency. The disclosed switching circuits canalso be used in equipment such as magnetic resonance circuits, medicalequipment circuits (e.g., ultrasound imaging or x-ray equipment).Another possible application is in telecommunications equipment wherethe antenna impedance should be matched to the transmitter outputimpedance. In such cases as transmitters, there are many applicationswhere the carrier frequency is switched under so-called hot switchingconditions. These switches are also used in handheld transmitters formilitary use, and in many other military applications using varioustransmitters and receivers.

Referring now to FIG. 11, a first switching circuit 1100 is shown. Thefirst switching circuit 1100 comprises a passive switch 1110 coupled toa first switch terminal A. The first switching circuit 1100 furthercomprises a driving switch M1 coupled in series with the passive switch1110 and a second switch terminal B, the driving switch M1 configured toturn the passive 1110 switch on and off. The first switching circuit1100 further comprises a power source 1160 configured to provide powerto the passive switch 1110 and the driving switch M1. The firstswitching circuit 1100 further comprises a monitoring circuit 1150configured to (1) receive an indication that a switching circuit voltageexceeds a predetermined amount and, in response, reduce the powerprovided to the driving switch M1; and (2) receive an indication that aswitching circuit current exceeds a predetermined amount and, inresponse, reduce the power provided to the driving switch M1. In otherembodiments, the monitoring circuit 1150 can be configured to performoperation (1) and not operation (2), or operation (2) and not operation(1). These components and operation will be described further below.

In the exemplified embodiment, the power source 1160 is a switched modepower supply (SMPS). Further, the exemplified switching circuit includesa gate driver 1102 operably coupled to (a) the power source 1160, (b) agate of the passive switch 1110, and (c) a gate of the driving switchM1. In other embodiments, other power sources, drivers, and/oramplifiers can be used as required by the components of the specificswitching circuit.

In the exemplified embodiment, the passive switch 1110 comprises aplurality of switches connected in series. Specifically, the passiveswitch 1110 comprises two high voltage and high current junction gatefield-effect transistors (JFETs) J1, J2. In other embodiments, thepassive switch can comprise any number of switches, and those switchescan be of various types. The exemplified JFETs of the passive switch1110 are connected in series to increase the voltage operation of theswitching circuit 1100. The passive switch 1110 can also compriseexternal body diodes.

The first switching circuit 1100 further comprises a driving switch M1coupled in series with the passive switch 1110 and a second switchterminal B. The driving switch M1 can be configured to turn the passiveswitch 1110 on and off. In the exemplified embodiment, the drivingswitch M1 comprises a high voltage and high currentmetal-oxide-semiconductor field-effect transistor (MOSFET). In otherembodiments, other switches can be used, such as a BJT, an IGBT, a GaNdevice, and even a relay (e.g., in special applications at low frequencyand high currents). Connecting the driving switch M1 and passive switch1110 in series allows the circuit to use an industry standard gatedriver integrated circuit (IC) with the driving switch M1.

The switching circuit 1100 can be configured such that JFETS J1, J2 arealways turned on, and thereby the JFETS J1, J2 can provide high voltageisolation 1170. Resistors R3 and R4 (which can have large values, e.g.,10 Megaohms) can connect the gate and drain of the JFETs J1, J2 to thesame potential and can force the JFETS to the on condition. The drainvoltages starting on terminal A in FIG. 11 can distribute equally on allthree transistors M1, J1, J2. In other embodiments, the switchingcircuit can include more than two JFETS in series, but the packaging ofthese devices must be such that the high voltage would not break down onthe junctions. A separate circuit that allows equal voltage distributionon the switches is omitted here, as such circuits (e.g., a snubbercircuit) are well known. These switches are typically heat sunk eitheron an air heat sink or water-cooled heat sink. All these switchingarrangements can also use GaN devices since in the ON state they conductcurrent in both directions. Driving GaN gates, however, requiresslightly different gate drivers.

The first switching circuit 1100 can also include an optocoupler 1106configured to electrically isolate the gate driver IC 1102 from themonitoring circuit 1150 (discussed below). The optocoupler 1106 candrive the gate driver IC 1102 on and off, thus also driving the firstswitching circuit 1100 on and off. Supply line +V can be around 10-15VDC and can be supplied by a switched-mode power supply (SMPS). DCisolation on the SMPS is expected to be greater than 4 kV for thisapplication, thus enabling the switching circuit 1100 to be used in ahigh voltage RF application. In the exemplified embodiment, the lowvoltage monitoring circuit 1150 is isolated from the drains and highvoltage switching voltage by at least 4 kV peak, if not more in otherapplications. For higher power applications on this switching circuit,the voltage separation can be raised even higher. The gate driver canhave the floating SMPS associated with it for that purpose.

The exemplified optocoupler 1106 and gate driver IC 1102 are notconnected to chassis ground GND but to a virtual ground. The main RFground (terminal B) is connected to the virtual ground. The virtualground can be designed similarly to the virtual ground shown in FIG. 1B,which is an aluminum plate with embedded tubes for water cooling.Mounted on the virtual ground can be variable capacitors or inductorsand a switching circuit for each. Thus, components of the switchingcircuit 1100 (e.g., power FETs) can be mounted on the virtual ground. Asdiscussed above, the virtual ground can be connected via a seriesnetwork directly to the chassis ground.

The exemplified virtual ground is floating, and it has an isolation ofthe systems grounds as large as the transformers and the optocouplerswill allow. For that reason, two or more such system switches can beconnected in parallel to increase the current switching capabilities.

The first switching circuit 1100 can be considered “smart” or“self-healing” as it has two automatic shut-off conditions—one forexcessive voltage and one for excessive current. Regarding excessivevoltage, in the exemplified embodiment, the switching circuit 1100 canbe switched off if a voltage on terminal A exceeds a predeterminedvalue. For example, the predetermined value can be 4,000 V.

An excessive voltage indication can be provided by a voltage sensingcircuit 1130, the voltage sensing circuit 1130 comprising (a) a firsttransformer T1 operably coupled to the second terminal B and (b) a firstdiode D1 operably coupled to the monitoring circuit 1150. The switchingcircuit 1100 can be designed such that, if the voltage at terminal A is4,000 V, then the voltage at diode D1 output (voltage sensing circuitoutput) is 4 VDC. The voltage at the diode D1 output can provideevidence of excessive voltage at terminal A or elsewhere in the circuit.

Diode D1 and the optocoupler 1106 can be connected to a monitoringcircuit 1150 for monitoring the switching circuit 1100 for excessivevoltage or current. In the exemplified embodiment, the monitoringcircuit 1150 can be configured to (1) receive an indication that aswitching circuit voltage exceeds a predetermined amount (excessivevoltage indication) and, in response, reduce the power provided to thedriving switch M1; and (2) receive an indication that a switchingcircuit current exceeds a predetermined amount (excessive currentindication) and, in response, reduce the power provided to the drivingswitch M1.

The monitoring circuit 1150 can include low-level logic components todetermine whether a shut-off condition is present. For example,regarding excessive voltage, a first comparator 1151 can be used toreceive the excessive voltage indication. The first comparator 1151 canreceive a threshold voltage V_(ref) on a first terminal and can be setto normally provide an output a logical value “1” (e.g., +4 V). Thefirst comparator 1151 can also receive a signal from the voltage sensingcircuit 1130. When the voltage at the voltage sensing circuit output(the monitor voltage) exceeds the V_(ref), the comparator can change itscondition to logical “0” (e.g., 0 V), thus causing the shut offcondition on gate 1153 for the control signal 1154.

The second shut off condition of the switching circuit 1100 can be basedon current. The switching circuit 1100 can include a current sensingcircuit 1140. The current sensing circuit 1140 can determine a currentpassing from a first switch terminal A to a second switch terminal B ofthe first switching circuit when the switching circuit is turned on bythe control signal 1154. That current can be determined by the RFimpedance load that the first switching circuit 1100 is switching on andoff. This current is similar to load current I_(L) in FIGS. 1A and 3-10.The current sensing circuit 1140 can include (a) a second transformer T2operably coupled to the second terminal B and (b) a second diode D2operably coupled to the monitoring circuit 1150. The current sensingcircuit 1140 can further include sense resistor R_(S). Sense resistorR_(S) is shown adjacent to terminal B in FIG. 11, and its surroundingcomponents are shown in more detail below as part of sensing circuit1140. Sense resistor R_(S) can provide a low resistance (e.g., 10-50mOhms). The circuitry that follows the sense resistor R_(S) can functionlike a high-speed operational amplifier with sufficient gain bandwidthproduct to replicate the RF current waveform. The gain in theoperational amplifier chain must be such that the switching circuit 1100can detect an appreciable RF voltage. In some cases, such as at lowerfrequencies and high currents, the switching circuit 1100 can use acurrent transformer in the source of driving switch M1 instead of asense resistor. If the inductance of the current transformer issufficiently small at the frequency of operation, it can be substitutedfor the R_(S) sense resistor.

The current in the source of the driving switch M1 can thus be detectedand received via the source resistor R_(S). The processing of thecurrent waveform is received by the source resistor R_(S) and theoperational amplifier circuitry that follows the detected waveform. Thesignature of the current in the source of the driving switch M1 can beused to see its amplitude. This current amplitude is detected and can beviewed on an oscilloscope to determine the shape and the frequencyspectrum, if necessary. Further on, after the output of the operationalamplifier A1, there is diode D2. The output of that diode D2 (used as adetector) is a DC voltage proportional to the RMS value of the sourcecurrent in the driving switch M1. This detected DC voltage waveform canbe sent to the second comparator 1152, with a reference voltage V_(ref)on the other terminal of the second comparator 1152. When the detectedvoltage, which is now proportional to the current in the source of thedriving switch M1, is larger than a predetermined value, this can beconsidered an excessive current indication. In this case, the referencevoltage can trip the second comparator 1152 output from its normallyhigh value (logic value “1”) to a low value (logic value “0”). Themonitoring circuit 1150 can thus detect that the current in the drivingswitch M1 has exceeded the predetermined value.

The monitoring circuit 1150 can include an AND logic gate 1153 to switchoff the gate driver IC 1102 when a shut off condition has occurred. Inthe exemplified embodiment, a three-input wide AND logic gate 1153 isutilized. The detector diodes D1, D2, as described above, rectify the RFwaveforms of detected RF voltage and current and covert them to DC sothat the signals can be used at the comparator terminals. Thecomparators 1151, 1152 receive signals from the detector diodes D1, D2.The comparer outputs can be logic signals that are received by the ANDlogic gate 1153. The AND logic gate 1153 can shut off the incomingcontrol signal 1154 if a high voltage or high current condition isindicated. In the exemplified embodiment, when a high voltage conditionis indicated the first comparator output is a logic value 0, and when ahigh voltage condition is indicated the second comparator output is alogic value 0. If the AND logic gate 1153 receives a logic value 0 fromeither comparator 1151, 1152, the AND logic gate 1153 can shut down thecontrol signal 1154 and thereby switch off the gate driver IC 1102. Inso doing, the monitoring circuit 1150 can ensure safe voltage andcurrent operation values. The set of safe operating values can be set bythe appropriate reference voltage values on the comparators 1151 and1152.

The foregoing circuitry can further signal an LED light to tell anoperator that the switching circuit 1100 has exceeded the prescribedvoltage or current value. The LED light can stay on as long as there isan excessive voltage or current condition. In other embodiments, theswitching circuit 1100 can utilize methods other than those describedabove for responding to excessive voltage or excessive currentconditions on the passive switch 1110 and the driving switch M1.

There are several advantages to the described self-healing switch. Othersolutions for shutting off such switches could include the use ofcontrol board algorithms. But such an approach is usually too slow toshut off the driving switch M1 off when the voltage becomes excessive.The exemplified analog hardware implementation is much faster. Thevoltage sensing circuit 1130 is designed such that its frequencybandwidth can be large enough to cover at least the third harmonic ofthe RF frequency that is being switched. For example, if the switchingcircuit 1100 is used in the standard ISM frequency of 13.56 MHz, thebandwidth of the voltage sensing circuit 1130 can be larger than 40.68MHz. This large frequency bandwidth also enables one to see the shape ofthe switched waveform during the switching conditions. That shape can beobserved at terminal A2 before the detector diode D1.

The exemplified switching circuit 1100 can prevent a catastrophic switchfailure in the case of high voltage transient returning back from thevacuum chamber via the matching network and on the RF switches. In highVSWR ratios, that condition could occur, for example, when setting upthe process recipe in the process chamber and other anomalies in thechamber.

The switching circuit 1100 can further include an intermediary switchM10 for enabling the passive switch 1110 to switch simultaneously withthe driving switch M1. In the exemplified embodiment, the intermediaryswitch M10 is a MOSFET, though in other embodiments other switches canbe used. In the exemplified embodiment, the intermediary switch M10 canallow the switching circuit 1100 to disconnect the gate of switch J1when switch M1 is in the OFF condition. This allows slightly larger highvoltage potentials on terminal A.

The exemplified intermediary switch M10 comprises a gate, a drain, and asource. The gate is operably coupled to the power source 1160. In thisembodiment, the gate of the intermediary switch M10 is connected tooutput of the gate driver 1102, though other designs can be used.Further, the drain of the intermediary switch M10 is operably coupled tothe passive switch 1110. In the exemplified embodiment, the drain of theintermediary switch M10 is connected to the gate of switch J1.

If the intermediary switch M10 were not in the gate of switch J1, thegate of switch J1 would be connected to the virtual ground or a bottomterminal B. The maximum voltage on the drain of the driving switch M1would be low in all situations. The drain of driving switch M1 wouldonly see the V_(GS) (gate-source voltage) voltage drop of switch J1. Inthis case, all the voltage drops from terminal A down to terminal Bwould be across the passive switch 1110—in the exemplified embodiment,switches J1, J2. Since the driving switch M1 is used only as a switchingdevice at relatively low voltages, the driving switch M1 would likely bean inexpensive and high speed device.

But in the exemplified embodiment, due to the intermediary switch M10,the gate of driving switch M1 and the gate of switch M10 can be switchedsimultaneously. As a result, the off condition of the series connecteddriving and passive switches M1, 1110 (e.g., vertically cascodedswitches M1, J1, J2) behave differently. Specifically, the high voltagefrom terminal A down to terminal B can be distributed more evenly.Instead of two transistors taking all the high voltage distribution,there can be three transistors distributing the high voltage—in theexemplified embodiment, switches J1, J2 and M1—since the gate of J1 isopened with the switching action of intermediary switch M10.

In this example, switch M1 should have a high voltage rating similar tothose of switches J1, J2. If the gate of switch J1 would be connecteddirectly to ground, the switching circuit could use switch M1 at arelatively low drain voltage. For lower switching voltages, such aconfiguration could be used. Switch M10 can be a low drain currentdevice with a high drain voltage breakdown that does not have to be heatsunk.

The switching circuit 1100 can further include voltage sense capacitorsC1, C2, C3 and C4. A typical design of such a high voltage probe is inthe ratio of 1000:1. The capacitors C1, C2, C3 can therefore be in theorder of 10 pF and at very high break down voltage. Capacitor C4 can beadjusted accordingly to read 1000:1 in voltage ratio.

Resistors R3, R4 can be included to keep the switches J1, J2 on.Further, these resistors R3, R4 can have very high values (e.g., 10 MegaOhms or more), and thus can prevent excessive bleed off of RF current inthe off condition.

In the exemplified embodiment, transformers T1, T2 can be pulsetransformers with a winding isolation in the order of greater than 4 kV.These transformers T1, T2 can detect small voltages in the order of fewmillivolts to a few volts. Both are used as detector transformers. Theycan be very small in size but, since they are pulse transformers, theymust have very large frequency bandwidth response. The bandwidth istypically determined so that the switching circuit 1100 can detect up tothe fifth harmonic of the RF signal being switched. For instance, if theswitching circuit was to switch a high voltage and high current waveformat a fundamental frequency of 10 MHz, the detecting transformers T1, T2can have their bandwidths up to 50 MHz to clearly replicate the drainvoltage after capacitive dividers C1, C2, C3 and C4 for the drainvoltage and the current transformer T2 that will replicate the sourcecurrent in switch M1.

FIG. 12 provides a schematic of a second switching circuit 1200. Thesecond switching circuit 1200 is very similar to the first switchingcircuit 1100 of FIG. 11, and therefore most of the discussion withregard to FIG. 11 applies to also to FIG. 12. Accordingly, comparablecomponents are shown, including gate driver 1202, optocoupler 1206,driving switch M1, voltage sensing circuit 1230, current sensing circuit1240, monitoring circuit 1250, first comparator 1251, second comparator1252, AND logic gate 1253, control signal 1254, and power source 1260.Voltage isolation 1270 is also shown.

The primary difference from first switching circuit 1100 is that thesecond switching circuit 1200 uses MOSFETs M1, M2, M3 in series, insteadof a combination of MOSFETs M1 and JFETs J1, J2. Thus, the passiveswitch 1210 uses MOSFETs M2, M3 instead of switches J1, J2. The MOSFETsM1, M2, M3 in the shown vertical position will typically have smallerinter-electrode capacitance. In particular, the MOSFET M3 will typicallyhave a smaller capacitance between the drain and gate and the drain tosource than a JFET for the same current. Since the capacitance will besmaller, the second switching circuit 1200 can operate at slightlyhigher frequencies. The MOSFETs M2, M3 must be kept on (as with switchesJFETS J1, J2), but each MOSFET has its own floating DC voltage to keepit in the on condition. Those floating gate-source voltages V_(GS2),V_(GS3) for M2 and M3 are provided by the power supply 1260. To providethe necessary floating voltages for MOSFETs M1, M2, and M3, the powersupply 1260 (an SMPS) is designed to have three separate secondarywindings from which to make three isolated DC voltages—V_(GS1) (thegate-source voltage of M1), V_(GS2) (the gate-source voltage of M2), andV_(GS3) (the gate-source voltage of M3). The MOSFET M1 is the masterswitch to turn on and off the whole vertical chain of MOSFETs M1, M2 andM3. In both the first switching circuit 1100 and the second switchingcircuit 1200, the clock input terminal to the power supply SMPS 1260 canbe from 100 kHz to 1 MHz in switch mode supplies. The power supply SMPSalso can use the external DC voltage +V_(DD) from an external DC supply.In one embodiment, the external DC voltage can be in the order of +15VDC. This DC supply can be used in the first switching circuit 1100 andthe second switching circuit 1200.

FIG. 13 illustrates parasitic capacitances on a switching circuit 1250.The switching circuit 1250 includes four MOSFET M51, M52, M53, M54connected vertically in series. In other embodiments, other switches canbe used. Switch terminals D and S are floating. First parasiticcapacitances C51-C62 are on each of the MOSFETs. Second parasiticcapacitances C_(SM1)-C_(SM4) are from the back body of the MOSFETs tothe mounting on the heat sink ground GND. Also shown is couplingparasitic capacitances C_(TX1)-C_(TX4) on the gate driver transformers.The parasitic capacitances present possible frequency limitations, andtherefore it can be advantageous to keep the parasitic capacitances aslow as practically possible. When designing switching circuits, it canbe helpful to first predict the parasitic capacitances and from themcalculate the maximum frequency of operation for the switching circuit.

FIG. 14 is a graph of a switched waveform 1260. The waveform 1260 is anexample waveform for one embodiment of the switching circuit. Thefrequency is 2 MHz. The RF voltage switched is 3000 V peak to peak. Theswitching off time is approximately 2 μs. Note the that the switch doesnot swing the full 3000 V due to inherited capacitance in the circuit.Such computer analysis of switching performance can be performed usingP-spice or similar software.

FIG. 15 is a simplified block diagram that provides a basic model forthe switching circuits described above. The switching circuit 1280includes terminals A and B. The outputs are floating. High power voltageand current terminals are electrically isolated from the logic drivenswitching conditions. Since the terminals A and B are floating, suchswitches can be connected in parallel to increase the currentcapability. Such switches can also be connected in series to achieveeven higher voltages. In one example, the switch can provide a highvoltage isolation 1170 of more than 5 kV.

There are several advantages to the switching circuits discussed herein.First, the switching circuits eliminate disadvantages associated withPIN diodes. PIN diodes require elaborate RF choke designs to support theoff condition and DC forward current for turning the PIN diode on. ThePIN diodes also require a high DC voltage to back bias the PIN diode tooff. Further, the disclosed switching circuits can be simpler toimplement. Further, the disclosed switching circuit designs allow forthe addition of further switches (e.g., more FETs) to handle even highervoltages when such switches are connected in series. Further, moreswitches can be connected in parallel to allow for higher currentcapability of the switch.

FIG. 16 is a schematic representation of a third switching circuit 1300.In this embodiment, the switching circuit 1300 utilizes gallium nitridehigh-electron mobility transistors (GaN HEMTs). The circuit 1300structure is similar to that of cascode or vertically connected FETs. Byvertical connection of the GaN HEMTs, the breakdown voltage on the totalswitch structure can be increased.

The switching circuit may be understood as comprising a first switch1310 and a second switch T35. The first switch 1310 is coupled to afirst switch terminal A, and comprises at least one GaN HEMT. In otherembodiments, other FETs can be used. In this embodiment, the firstswitch 1310 comprises four GaN HEMTs T31-T34 connected in series. Inother embodiments, other numbers of transistors can be used. Using moretransistors will increase the breakdown voltage of the overall switch,thus allowing higher voltage on the switch in the OFF condition.

The second switch T5 is coupled in series with the first switch 1310 anda second switch terminal B. The second switch comprises a GaN HEMT,though other FETs can be used. The second switch is configured to drivethe first switch ON and OFF.

The switching circuit 1300 further comprises at least one isolated powersource configured to provide isolated power to the first switch and thesecond switch. In this embodiment, the power sources for the switchingcircuit 1300 include transformers TX1, TX2 and a DC power supply (notshown).

Second switch T5 is externally driven by a control signal from a gatedriver 1302B. In this embodiment, the gate driver 1302B must also behigh-voltage isolated from the low-level circuitry. For this purpose, acommercially available gate driver can be used with the GaN device(e.g., the SI8271GB-IS isolator from Silicon Labs). Such a gate drivercan withstand high-voltage isolation, for example, up to 5 kVDC. Thegate driver 1302B can receive DC power from a commercially available DCpower supply (e.g., PES1-S5-S9-M from CUI, Inc.). This DC power supplywould be isolated and, in this embodiment, can be provided at VDD of thegate driver 1302B. The switching circuit 1300 can be designed such thatall transistors from T31 to T35 are turned ON and OFF at precisely thesame time to achieve low switching losses.

In the exemplified embodiment, there are isolated gate drivers 1302A oneach of the gates of the passive transistors T31 to T34. The gatedrivers 1302A include transformers TX1, TX2. In other embodiments, otherconfigurations of power sources can be used to drive the gates. In thisembodiment, the gate drivers 1302A are specially designed to minimizecost. According to the exemplified embodiment, the ON/OFF pulses arerectified by diodes D31-D34 and smoothed out by capacitors C31-C34.Resistors RG1-RG4 are static discharge resistors and can have a value,for example, of 10 kohms. Similar to the gate driver 1302B of drivingswitch T35, the gate drivers 1302A can receive DC power from a DC powersupply (not shown). This DC supply must also be isolated to at least 5kV in the exemplified embodiment. These DC low power supplies arecommercially available (e.g., PES1-S5-S9-M from CUI, Inc.), andtherefore are not discussed in detail. When the gate drivers provide therequisite voltage (e.g., 6V) to the gate of the transistor, thetransistor is turned on.

For better distribution of voltages in the OFF condition on the GaNHEMTs, resistors R31-R34 (e.g., 10 Mega Ohms) can be included inparallel to each drain to source. Further, it is noted that theswitching circuit 1300 (and the other switching circuits discussedherein) can further include a monitoring circuit similar to thatdiscussed above.

One advantage of the disclosed circuit 1300 is that, by using GaN HEMTs,when the switch is fully ON at RF frequencies, RF AC current can flow inthe positive and negative directions. The switching circuit can passfull RF current in both directions. The GaN HEMTs also work well at highfrequencies. It is noted that GaN HEMTs can also be used in the otherswitching circuits discussed in this disclosure.

FIG. 17 is a schematic representation of a fourth switching circuit1400. This switching circuit 1400 is designed to sustain high voltageswings in the OFF condition. In one embodiment, silicon carbide (SiC)FETs having a breakdown voltage of 1200V are utilized for devicesM41-M44. Such a switch can swing the AC RF voltage in the OFF conditionto at least 4,000V peak to peak.

In the exemplified embodiment, the configuration of FETs M41-M44 is suchthat, when the switch is in the OFF condition, a full RF voltage ofapproximately 4 kV peak-to-peak can be applied between terminals A and Band not conduct any RF current through the FETs. When in the OFFcondition, during the positive swing on the terminal A, the body diodesD41 and D42 of FETs M41 and M42 will keep the positive voltage on A inthe OFF condition. Likewise, when the negative RF voltage swings onterminal A (in the OFF condition), the body diodes D43 and D44 will notallow the RF current conduction to ground. Some leakage current mayoccur through the bleeder resistors R41-R44 but will be minor.

In the exemplified embodiment, the bleeder resistors R41-R44 are 10 MegaOhms and are in parallel to all drain to source terminals on the FETsM41-M44. These resistors R41-R44 can keep equal voltage distribution onthe drain to source on all devices when OFF. This will also keep all thedrain to source voltages symmetrical. It is understood that the valuesdiscussed herein are examples and other appropriate values may be used.

The exemplified switching circuit may be understood as comprising afirst switch 1410 and a second switch 1420. The first switch cancomprise a single transistor M42 or more transistors M41. The one ormore transistors M41, M42 are operably coupled in series with a firstterminal A. Each of the one or more transistors M41, M42 has acorresponding diode D41, D42. Further, a drain of each of the one ormore transistors is operably coupled to a cathode of the correspondingdiode.

The second switch 1420 can comprise a single transistor M43 or moretransistors M44. The one or more transistors M43, M44 can be operablycoupled in series with a second terminal B. Each of the one or moretransistors can have a corresponding diode D43, D44. A drain of each ofthe one or more transistors can be coupled to a cathode of thecorresponding diode. Further, a source of the one or more transistors ofthe first switch 1410 can be operably coupled to a source of the one ormore transistors of the second switch 1420. Thus, in the exemplifiedembodiment, the source of M42 is coupled to the source of M43. A drainof transistor M41 is coupled to terminal A, and a drain of transistorM44 is coupled to terminal B. The transistors of each switch 1410, 1420can be coupled source-to-drain. Each of the transistors can furtherinclude a corresponding resistor R41-R44 operably coupled in parallel tothe corresponding diode. In other embodiments, M41 and M44 can beeliminated, or additional transistors can be added, provided the generalfeatures of and relationship between the first and second switch (asdescribed in the independent claims) are maintained.

Gate drivers GD1, GD2, GD4 are isolated and can utilize commerciallyavailable gate drivers. These gate drivers are typically designed tosustain at least 5 kV peak voltages. These gate drivers also require afloating DC/DC supplies 1430. The floating DC supplies are alsoisolated. Further, the DC/DC supplies 1430 are also commerciallyavailable and thus will not be separately discussed.

The cascoded FETs can be increased to a higher number such as six ormore devices connected vertically. In that case the higher RF AC voltageswing in the OFF case will be achieved. For that connection, the gatedrivers must sustain higher breakdown voltages. Also, the floating orisolated DC/DC supplies must sustain higher isolation voltages. It isnoted that the different potential characteristics of gate drivers andDC power supplies discussed herein can also apply to the gate driversand DC power supplies of this switching circuit 1400.

The switching circuits 1300 and 1400 have several advantages. The risetime of the ON/OFF condition can be achieved in the order tens ofnanoseconds or less. They also enable low capacitance to ground andminimize leakage.

Driver Circuit for Diode

As discussed above, in industries such as semiconductor manufacturing,there is need for driver circuits that provide faster switching toenable faster impedance matching. FIG. 18 provides a switching circuit 3comprising a driver circuit 4 for a PIN diode 5 according to oneembodiment. The exemplified driver circuit 4 has ON/OFF drivingcapabilities for a high voltage and high current PIN diode 5 atfrequencies from 400 kHz to 100 MHz. In other embodiments, the conceptsdiscussed herein can be applied to other switches (including NIP diodes)and in other suitable applications.

The exemplified driver circuit 4 is used to control PIN diodes 5 of anEVC used for the digital operation of the matching network. Theinvention is not so limited, however, as the switching circuit and itsdriver circuit can be used to switch other types of switches in othertypes of systems.

In the exemplified embodiment, the driver circuit 4 includes a firstswitch 11 and a second switch 12 coupled in series, where a drain of thefirst switch is coupled to a source of the second switch. Further, thefirst and second switches 11, 12 are metal-oxide semiconductorfield-effect transistors (MOSFETs). In other embodiments, however, thefirst and second switches can be other types of transistors or switches.In the exemplified embodiment, both the first switch 11 and the secondswitch 12 have a breakdown voltage of at least 1,700 VDC. When the firstswitch 11 is turned ON, the PIN diode 5 is biased in the forward currentdirection and thus turned ON. The forward ON current can be adjusted toan optimal set point to minimize the losses of the PIN diode 5. For thePIN diode 5 current adjustment, sense resistor 15 can be used to providefeedback to the controller. When the PIN diode 5 is turned ON, the RFcapacitor 7 is connected to ground. In this embodiment, the capacitor 7is one of many in an array of capacitors of an EVC. These individualcapacitors can be switched ON or OFF according to a predeterminedalgorithmic sequence to vary the EVC's total capacitance.

The second switch 12 is used to switch OFF the PIN diode 5. In thisembodiment, the transistor 12 has a sufficiently large breakdown voltageat DC to switch a high DC voltage, such as 1,600 VDC, to the cathode ofPIN diode 5. When second switch 12 is turned ON, as much as the full1,600 VDC can be applied to the cathode of the PIN diode. The PIN diode5 is then reversed biased and thus the PIN diode 5 is turned OFF.

The driver circuit 4 is controlled by drivers 21, 22, and 23.Specifically, first gate driver 21 is operably coupled to first switch11, and second gate driver 22 is operably coupled to second switch 12.Third gate driver 23 is operably coupled to the first and second gatedrivers 21, 22. In the exemplified embodiment, the third gate driver 23is a half bridge gate driver, but the invention is not so limited. Thethird gate driver 23 is configured to provide a first signal to thefirst gate driver, and a second signal to the second gate driver. Thefirst and second signals substantially asynchronously drive the firstand second gate drivers on and off. Note that the gate drivers discussedherein can comprise an integrated circuit or a discrete circuit. Inother embodiments, any of the gate drivers can be replaced with othertypes drivers for driving switches or other drivers.

FIG. 19 provides a timing diagram 80 of the sequence of switching thefirst and second switches 11, 12 of the driver circuit 4 of FIG. 18.This waveform is generated in the third gate driver 23 of FIG. 18. As isshown, there can be dead times 81 or time delays between (a) the thirdgate driver 23 driving the first gate driver 21 ON and the second gatedriver 22 OFF, or (b) the third gate driver 23 driving the second gatedriver 22 ON and the first gate driver 21 OFF. Accordingly, there canalso be dead times between switching the first switch 21 ON and thesecond switch 22 OFF, or vice versa. These dead times 81 can beexceptions to the otherwise asynchronous switching of the first andsecond switches 21, 22. The duration of these dead times 81 can beadjusted by varying the resistance of potentiometer 23R on the gatedrive 23 in FIG. 18. The dead time 81 can ensure that the energy-storinginductor 9L of the filter circuit 9 does not create a large voltagespike. The dead time 81 can be varied based on the size of the inductor9L.

Returning to FIG. 18, as discussed above, in the exemplified embodiment,each switch 11, 12 is driven by its own gate driver 21, 22. The gatedrivers are isolated between the logic input driving node and the outputnode. The drivers can have a breakdown voltage in excess of 3 kV_(rms)at the frequency of operation. Separate floating power supplies can feedthe gate drivers 21, 22. The gate drivers 21, 22 can also be excited bythe third gate driver 23 that supplies the voltage waveform as depictedin FIG. 19. The exemplified gate driver 23 is on the low side of thedriver input and does not need the floating low voltage power supply. Asdiscussed above, the third gate driver 23 has a provision to adjust thedead time between ON and OFF states of switching the first and secondswitches 11, 12. This dead time is adjusted, at a moment, by thepotentiometer 23R connected to a ground.

In the exemplified embodiment, between the switching node Ni and thebottom of the switched capacitor 7, connected to the cathode of the PINdiode 5, is an LC or parallel-tuned circuit 9 at the operatingfrequency. The LC circuit 9 comprises an inductor 9L and a capacitor 9Ccoupled in parallel. The LC circuit 9 is coupled at a first end betweenthe first switch 11 and the second switch 12, and coupled at a secondend to the diode 5.

The exemplified LC circuit 9 is utilized when the PIN diode 5 is in theOFF state to maintain very high impedance at the bottom of switchedcapacitor 7. This LC circuit 9 assures minimum RF current leakage fromthe switched capacitor 7 to ground in the OFF state. Also, in parallelto this inductance is the parasitic capacitance of the PIN diode 5 aswell as packaging capacitance. All of it can be parallel tuned to makesure the impedance is the highest at the OFF condition of the PIN diode5. The invention, however, is not limited to the use of LC orparallel-tuned. In other embodiments, other filter circuits, such as alow pass filter, can be utilized.

When the PIN diode 5 is in the OFF state, the total capacitance of theswitch in OFF condition should be low, that is, at least about ten timeslower that the value of the switched capacitance 7. For instance, if theswitched capacitance 7 has a value of 100 pF, it is desirable that atotal capacitance from the lower side of capacitor 7 to ground includingthe PIN diode 5 OFF capacitance be less than 10 pF.

The exemplified filter traps 9 are put individually on each PIN diode 5in the capacitive array. Such filter traps can be challenging to designwhen operating at higher frequencies (>40 MHz) with high RF voltageswings on the RF bus. For the PIN diode 5 ON case, a forward DC currentadjustment on each PIN diode 5 is measured by the low value of the senseresistor 15 in series with the negative supply. The current sensevoltage drop on sense resistor 15 is amplified and processed by anoperational amplifier (op-amp) 17. The output voltage of the op-amp 17,now proportional to the current, is optically coupled by a linearopto-coupler 18 to an output node 19, the output node 19 having anoutput sense voltage, V_(s). The control board then processes the outputsense voltage, V_(s). The control board (not shown here) can measure thecurrent in each PIN diode 5 to make sure all PIN diodes 5 havesufficient DC current bias available for processing the RF current whenthe PIN diode 5 is ON. In the exemplified embodiment, a bias current of0.5 A gives sufficiently low ON resistance in the PIN diode 5.

The sense voltage V_(s) can also be used to automatically set the DCcurrents in each PIN diode 5 in an array of diodes. A typicalapplication may use as many as 48 PIN diodes to switch the RFcapacitances. The automatic setting of the DC bias current in PIN diode5 may require the algorithmic voltage adjustment of the negative DCsupply. These automatic settings can be embedded in the control board.

As discussed above, the switching circuits discussed above can be usedas part of a matching network, such as the matching networks discussedin FIGS. 1A-10 discussed herein. These matching networks are shownwithin large RF semiconductor manufacturing systems that include an RFgenerator (or RF source) and a plasma chamber. The switching circuitscan be used with n-configuration, L-configuration, and T-configurationmatching networks, as well as other types of matching networkconfigurations. Further, the driver circuit is not limited to PINdiodes, as it can be used for other types of diodes or switches.Further, the driver circuit can be used in applications unrelated tosemiconductor manufacturing. Further, while the above embodimentsdiscuss providing a variable capacitance, the switching circuit cansimilarly be utilized to provide a variable inductance.

The switching circuit discussed herein provides several advantages. Thecircuit can enable high RF voltage and high RF current switching usingthe PIN diodes. Further, the switching circuit can use only two highvoltage switches (e.g., MOSFETs) to achieve a desired switchingcondition for high voltage and high current RF PIN diodes. Further, theswitching circuit enables very high switching speeds for turning ON/OFFa PIN diode. Further, the switching circuit enables low switching losseson the PIN diode, and very low thermal loss on the PIN diode since theDC current through the PIN diode is adjusted to have the lowest Ronresistance, typically in the 50 mΩ range. Further, the galvanicisolation between the high voltage switched side and the input logicdrive is can be large (e.g., in excess of 3 kV_(rms)). Further, as shownin FIG. 18, each switching circuit can be laid out in column, allowingvery compact design with RF and galvanic isolation among the multitudeof switches. Further, the switching circuit enables the automaticsetting of forward bias currents in each individual PIN diode, and theautomatic shutdown of the PIN diode switch if it draws an excess of thecurrent, a self-healing process of the RF switch.

Parasitic Capacitance Compensation Circuit

As discussed above, an EVC typically has several discrete capacitors,each having a corresponding switch. In the OFF state, parasiticcapacitance on the switches can be significant. There is need for acompensation circuit that can tune out such parasitic capacitances.

FIG. 20 is a schematic of a system having parasitic capacitancecompensation circuits for discrete capacitors of an EVC according to oneembodiment. EVC C comprises an array of discrete capacitors. In theexemplified embodiment, EVC C includes discrete capacitors C1, C2, andC3. Each of these discrete capacitors C1, C2, and C3 can be switched outby a respective switch S1, S2, S3. For example, switch S1 switchesdiscrete capacitor C1 in and out. The total capacitance of EVC C(referred to herein as “C”) can be altered by altering the discretecapacitors switched in or out. In the exemplified embodiment, threediscrete capacitors are shown, though in other embodiments other numbersof discrete capacitors can be utilized.

Each switch S1, S2, S3 is operably coupled to a parasitic capacitancecompensation circuit PCC1, PCC2, PCC3, which compensates for parasiticcapacitance in an RF switch, and which will be discussed in furtherdetail below. The circuits of FIG. 20 are simplified and divided intosections to more easily describe the function of the system.

Further, EVC C is coupled to an RF power source V1. The EVC C and RFsource V1 can be any of the EVCs or RF sources discussed herein,including those forming part of a matching network. Such matchingnetworks can have any type of the matching network typology, such as api-, or L-, or T-type. While the PCCC discussed below is used incompensating for parasitic capacitance in a matching network utilizingEVCs, the invention is not so limited, as it may be used for other typesof switches having parasitic capacitances.

FIG. 20 further shows details of switch S1 and its correspondingcompensation circuit PCCC1. Switch S1 switches discrete capacitor C1 inand out. Compensation circuit PCCC1 compensates for the parasiticcapacitance of switch S1. To simplify FIG. 20, the details of theadditional switches S2, S3 and compensation circuits PCCC2, PCCC3 arenot shown, though similar circuitry can be used. Terminal A and terminalB represent to two terminals of switch S1. They can also represent theterminals of the compensation circuit PCCC1, which couple to the twoterminals of the switch S1.

For switch S1, FIG. 20 identifies all the possible parasiticcapacitances between terminals A and B of the switch S1. Some of thecapacitances on the switch S1 have fixed values and some are variablewith the RF peak voltage impressed between terminals A and B. Outputcapacitances Co1, Co2 on the RF switch vary with the peak voltagevariation on the terminals A, B. FIG. 20 further shows gate-to-drainparasitic capacitances Cgd1, Cgd2, and gate-to-source parasiticcapacitances Cgs1, Cgs2.

FIG. 21 is a graph showing output capacitance variation for a typicalSiC high-voltage MOSFET that may be used as a switch according to oneembodiment. In other embodiments, other types of switches, includingother types of FETs, may be used. FIG. 21 specifically shows acapacitance variation for an input capacitance (curve Ciss), an outputcapacitance (curve Coss), and reverse transfer capacitance (Crss).Output capacitance Coss corresponds with output capacitance Co1 oroutput capacitance Co2 of FIG. 20. The abbreviations Co and Coss areused interchangeably herein.

A typical application of such an RF FET switch is in a digital matchingnetwork where the output voltage on the terminals A, B is about 600Vpeak (3 kW and 13.56 MHz). At this power level and peak voltage, FIG. 21indicates the output capacitance Coss is 60 pF. When the voltage on theswitch terminals A, B falls down to 160 V peak (about 150 W on thematching network input) the capacitance of the terminals A, B as readfrom FIG. 21 as Coss rises to 120 pF. Thus, at a most typical peakvoltage for the switch of 600V (3 kW), the output capacitance of theswitch in the OFF state is 60 pF. When the peak voltage drops to 160V(150 W), the output capacitance of the switch in the OFF state doublesto 120 pF. The most typical peak voltage can vary depending on theapplication in which the switch and corresponding compensation are used.The most typical peak voltage can be any peak voltage most frequentlypresent at the switch or compensation terminals when the system isoperating.

In the exemplified embodiment, the parasitic capacitance of the switchin the OFF state is compensated for by tuning out at two discretepoints, though in other embodiments additional points may be used. WhenRF power is varied from 150 W to 3 kW (a factor 20) the capacitance inthe OFF state changes only from 120 pF to 60 pF, that is, only by afactor of 2. As can be seen from FIG. 21, the capacitance change with Vpeak on the drain is a very shallow curve. Selection of a two-point tuneout circuit may therefore be adequate. While adding more tuning pointsis an option, fewer tuning points can decrease costs and parts.

In the exemplified embodiment, the tuning out of parasitic capacitanceis achieved as follows. The total parasitic capacitance Cpar on the RFswitch shown in FIG. 20 is:Cpar=Cgd+Co+Cpg

Thus, the total parasitic capacitance takes into consideration thegate-to-drain capacitance Cgd, the output capacitance Co, and packagingcapacitance Cpg. Packaging capacitance Cpg can be caused by, forexample, when a FET is put down on a PCB and soldered down. Theadditional capacitance to ground caused by the soldered pin or anattached pin on the drain of the FET can create such a packagingcapacitance Cpg.

In the exemplified embodiment of RF switch S1, transistors T1 and T2 areused to control RF current flow in both directions through the switch S1when the switch S1 is either ON or OFF. In other embodiments, otherswitching arrangements and components can be used. For example, in theexemplified embodiment, transistors T1 and T2 are FETs, but other typesof switches may be used. In this embodiment, Cgd can be based on Cgd1and Cgd2, Co can be based on Co1 and Co2, and Cpg can be based on Cpg1and Cpg2 as shown in FIG. 20.

In one example, when the switch S1 is OFF, Cgd is 3 pF, Co1 is 60 pF,and Cpg is 10 pF, and therefore a total parasitic capacitance is Cpar is73 pF. One can then calculate the parallel resonant condition for thefirst case when Co is 60 pF and the peak voltage on the switch S1 is600V or higher. Parallel resonance will provide minimum current leakagethrough capacitor C1.

The parasitic capacitance Cpar will be resonated out in a parallelcircuit configuration. First inductor L1 is operably coupled betweenfirst terminal A and second terminal B, the first inductor causing afirst inductance between the first and second terminals A, B. Theparasitic capacitance Cpar resonates with discrete inductance L1 asshown in FIG. 20. When they resonate, the parallel impedance of thecircuit will be the highest and the leakage current through discretecapacitor C1 will be the lowest. At this moment when the switch is OFFand a high voltage of 600V peak or higher is impressed on capacitor C1,the current should be the minimum through the switch.

${L\; 1} = \frac{1}{\omega^{2}{Cpar}}$

In this example, f is 13.56 MHz, and ω=2πf. Thus, L1 is 1.9 uH.

This inductance value is used for inductor L1 of FIG. 20. When thevoltage on the terminals A, B decreases to at or below 160V peak, theparasitic capacitance increases due to the increase of outputcapacitance of the FET T1, T2. At that drain potential the Cocapacitance increases to 120 pF as discussed above.

At this moment where the parasitic capacitance has increased by a factor2, the compensation circuit PCCC1 adjusts the tuning conditions. The newparasitic capacitance Cpar is:Cpar=Cgd+Co+CpgCpar=5+120+10=135 pF

The total equivalent inductance for this condition is recalculated at13.56 MHz as:

${L\; 2^{\prime}} = \frac{1}{\omega^{2}{Cpar}}$ L 2^(′) = 1  uH

The inductance of the second inductor L2, as shown in FIG. 20, is beingswitched on by the JFET T3 (or other switch) in series with secondinductor L2 when the voltage on the terminals A, B falls at or belowabout 160 V peak. Therefore, the inductance L2 must be recalculatedsince the parallel combination of L1 and L2 must be equal to L2 ^(/)=1uH. Hence, in this case inductor L2 has a value of 2.1 uH.

As shown, the second inductor L2 is operably coupled between the firstand second terminals A, B and parallel to first inductor L1. The secondinductor L2 causes a second inductance between the first and secondterminals of the switch when the second inductor L2 is switched in. Thesecond inductor L2 is switched in when the peak voltage on the first andsecond terminals falls below a first voltage, which is 160V in thisembodiment. The first inductance L1 tunes out substantially all of aparasitic capacitance of the switch when the switch is OFF and the peakvoltage is above the first voltage. The first and second inductances L1,L2, collectively tune out substantially all of the parasitic capacitanceof the switch when the switch is OFF and the peak voltage is below thefirst voltage. As used herein, the term substantially all means at leasteight five percent (85%).

FIG. 22 is a graph showing a power curve for a typical SiC high-voltageMOSFET according to the exemplified embodiment. Curve 310 shows amaximum power dissipation allowed on the drain-to-source junction when50 W dissipation is allowed at the device's case temperature of 95° C.

FIG. 23 is a graph of a maximum power dissipation of a typical SiChigh-voltage MOSFET according to the exemplified embodiment. It is notedthat the graphs of FIGS. 21-23 are adapted from graphs from CREE Corp.FIG. 23 shows the power dissipation of 50 W at the case temperature of95° C. This allows the calculation of the maximum allowable draincurrent versus the drain-to-source voltage. Curve 310 of FIG. 22 showsthe drain-source voltage versus the drain-source current at a maximumpower dissipation of 50 W and 95° C. case temperature.

TABLE 2 Max Drain V_(DS) [V]DC Current [mA or RMS Peak [V] RMS] MOSFET800 1,131 63 OFF 600 849 83 OFF 400 567 125 OFF 200 283 250 OFF 100 141500 OFF 50 71 1 A Getting ON 25 35 2 A Getting ON 10 14.4 5 A ON 5 7.110 A  ON

Returning to FIG. 20, in the exemplified embodiment, capacitors Cc1 andCc2 are used to sample the RF voltage on the switch at terminal A. Thesampled voltage is processed via the transformer L3 that allows takingsome energy out of the capacitive divider to keep the JFET sub-switch T3in the OFF condition until the voltage on terminal A falls below about160V. At that moment, the detected DC voltage on diode D1 falls down toless than minus 1V, in which case the JFET turns ON. This causesinductor L2 to switch ON, which resonates with the new higher parasiticcapacitance (120 pF), and thus tunes out the new larger capacitance.

The described parasitic capacitance compensation circuit PCCC1 can tuneout parasitic capacitances in all solid state switches (such as MOSFETS,GaN FETS, SiC FETS, BJT, and IGBTs and others) and any other switchcircuits that suffer from excessive capacitance. Such a circuit canallow switches, including ordinary RF FETS with large outputcapacitances, to operate at higher frequencies. The compensation circuitcan become an integral part of a solid state switch for an EVC. In apreferred embodiment, the compensation circuit is tightly packagedelectrically and thermally to the designed switch. Thermal issues on theRF switch and the compensation circuit can be solved simultaneously,since after the compensation circuit is invoked, the RF switch andcompensation circuit become one unit.

The compensation circuit can also be applied to PIN diodes and evenhigh-power electronic transmitter tubes used in HF heating applications,etc. In some cases, the compensation circuit can be added externally toa solid state relay device and improve the solid state relay conditionwhen in the OFF state. The invention not limited to specified types ofswitches, transistors, or other components shown in the embodimentsdiscussed above.

As discussed, the switching circuits and compensation circuits discussedabove can be used as part of a matching network, such as the matchingnetworks discussed in FIGS. 1A-10 discussed herein. These matchingnetworks are shown within large RF semiconductor manufacturing systemsthat include an RF source and a plasma chamber. The switching circuitscan be used with π-configuration, L-configuration, and T-configurationmatching networks, as well as other types of matching networkconfigurations. In a method for manufacturing a semiconductor, asubstrate is placed in the plasma chamber, the plasma chamber configuredto deposit a material layer onto the substrate or etch a material layerfrom the substrate. The plasma in the plasma chamber is energized bycoupling RF power from an RF source into the plasma chamber to perform adeposition or etching. While energizing the plasma, the matching networkbetween the RF source and plasma chamber can carry out an impedancematch. The matching network can include an EVC whose discrete capacitorsuse a parasitic capacitance compensation circuit similar to thatdiscussed herein. Further, the parasitic capacitance compensationcircuit can form part of an EVC of a matching network where the matchingnetwork forms part of a semiconductor processing tool. The semiconductorprocessing tool can comprise a plasma chamber and an impedance matchingnetwork, such as any of the plasma chambers and impedance matchingnetworks described herein (e.g., matching network 100 and load 120 ofFIG. 1A).

Switching Circuit Cancelling Parasitic Capacitance

As discussed above, an EVC typically has several discrete capacitors,each having a corresponding switch. There is need for a switch that cancancel the effects of parasitic capacitances on the switch. Thefollowing provides embodiments for a circuit architecture of asingle-pole, high-power RF switch. Any active switching element can beused, such as BJTs, IGBTs, JFETs, MOSFETs, HEMTs, etc. Silicon carbide(SiC) and gallium nitride (GaN) devices have good properties forhigh-power RF switches; however, more traditional silicon devices mayalso be used.

FET transistors can be used as a switching element at RF frequencies.These RF switches are typically used in matching networks at ISMfrequencies and at power levels from 500 W to 5 kW. The predominantapplications are in the semiconductor industry when driving a plasmachamber. The switch circuit architecture described herein will allow thedesign of new matching networks at power levels up to 10 kW using thepresent technology of RF power transistors.

FIG. 24 is a simplified schematic of a system 554 utilizingelectronically variable capacitors 552, 555 according to one embodiment.The exemplified system 554 comprises an RF source 510 and asemiconductor processing tool 553. The semiconductor processing toolincludes a matching network 550 and a plasma chamber 520. In thissimplified schematic, the matching network 550 comprises a firstelectronically variable capacitor (EVC) 552 and a second EVC 555arranged in a pi-configuration, though it is understood that otherconfigurations (such as those L, and T-type configurations discussedherein) can be utilized. The matching network 550 has an RF input 512for operably coupling to the RF source 510, and an RF output 513 foroperably coupling to a plasma chamber 520 or other load.

The EVC 552 comprises an RF bus 556 having discrete capacitors 541, 542,543, 545 and corresponding switches 531, 532, 533, 535. Each switch 531has a first terminal 521 operably coupled to the corresponding discretecapacitor 541, and a separate second terminal 522. In this embodiment,the second terminals 522 are operably coupled to each other and to areference voltage (ground). The switches 531, 532, 533, 535 areconfigured to switch in and out the discrete capacitors 541, 542, 543,545 to alter a total capacitance of the EVC 552. This allows for aprecise and fast match with no moving parts. The exemplified switchesare designed to be floating devices to some prescribed high-voltage (HV)level.

In other embodiments, other numbers of capacitors can be used. In yetother embodiments, other variable reactance elements comprising one ormore reactance elements can be used, such as an electronically variableinductor comprising one or more inductors. Further, it is noted that inother systems, the RF source 510, matching network, 550, and/or plasmachamber 520 can have alternative characteristics such as those discussedin the other embodiments provided herein.

FIG. 25 is a schematic of high-power RF switch 531 for discretecapacitor 541 of electronically variable capacitor 552 of FIG. 24. Ascan be seen, each switch 531 includes a switching circuit 546 and atuning circuit 547 coupled in parallel. The tuning circuit 547 can usedto tune out a cumulative parasitic capacitance of the switching circuit546. The exemplified tuning circuit 547 includes a tuning inductor 579and a tuning capacitor 578 coupled in series, though the invention isnot so limited. For example, in another embodiments, the tuning circuitcan omit the tuning capacitor, as the tuning capacitor is used mainly asa bypass capacitor to prevent any direct current that could develop inthe circuit from leaking through the tuning inductor 579 and potentiallydetuning the circuit. If used, the tuning capacitor 578 will generallybe large to look low impedance at the operating frequency. The tuningcircuit will be discussed in much greater detail below.

The exemplified switching circuit 546 is coupled between the firstterminal 521 and the second terminal 522. The switching circuit 546includes a switching transistor 570 having a transistor first terminal571, a transistor second terminal 572, and a transistor third terminal573. In the exemplified embodiment, the switching transistor 570 is aSiC FET, and the transistor first terminal 571 is a gate, the transistorsecond terminal 572 is a drain, and the transistor third terminal 573 isa source. But as discussed above, this transistor for carrying out theswitching action is not so limited and could be any power FET or BJTdevice. The FET is the dominant RF power device at this moment in powersemiconductor development. The switching of the switching transistor 570ON and OFF causes its corresponding discrete capacitor 541 to beswitched in and out.

In the exemplified embodiment, the switching transistor 570 is imbeddedin a cradle of four HV switching diodes 581, 582, 583, 584. Firstterminal 521 is connected to one electrode of an RF capacitor and secondterminal 522 is connected to ground, though the invention is not solimited. First diode 581 has an anode operably coupled to the secondterminal 522 and a cathode operably coupled to a drain 572 of thetransistor 570. Second diode 582 has an anode operably coupled to thefirst terminal 521 and a cathode operably coupled to the drain 572 ofthe transistor 570 and the cathode of the first diode 581. Third diode583 has a cathode operably coupled to the first terminal and an anodeoperably coupled to the source 573 of the transistor 570. Fourth diode584 has a cathode operably coupled to the second terminal and an anodeoperably coupled to the source 573 of the transistor 570 and the anodeof the third diode 583. Note that the invention is not limited to theuse of diodes as switching devices. As will be discussed further below,the diodes can be replaced with other switching devices, such astransistors. In one embodiment, the switching devices are insulated-gatebipolar transistors (IGBTs).

The switching transistor 570 is turned ON and OFF by its HV isolatedgate driver circuit 575. The gate 571 and source 573 of the transistor570 are coupled to the driver circuit 575. In the exemplifiedembodiment, the gate driver 575 for transistor 570 is floating and thusis isolated to at least >5 kV peak voltages. The source and drainterminals 573, 572 of transistor 570 will be floating on high DC voltagewhen device is switched OFF.

The exemplified transistor 570 and diodes 581, 582, 583, 584 arethermally connected to a heat sink via a good thermal conductor that isat the same time electrically isolated. Alumina or aluminum nitridesubstrates and silicone-based thermal greases are typically used forthermal management. It is important to take the heat out of the powersemiconductor devices when they are switching high currents. Theelectrical properties of these devices are dependent on temperature.

There will be a parasitic capacitance from source and drain oftransistor 570, first terminal 521 to ground, and second terminal 522 toground. This is shown in the subsequent figures. Switching diodes shouldbe fast switching diodes with breakdown voltages equal to or higher thanthe switching transistor 570. These diodes should be chosen to havetheir parasitic capacitances as low as possible.

Condition When RF Switch is ON

FIG. 26 is a schematic representing the high-power RF switch 531 of FIG.25 in the ON state. The switching diodes 581-584 have their parasiticcapacitances shown and represented by their notations as C_(D1), C_(D2),C_(D3), and C_(D4). Typical values of parasitic capacitance of thehigh-voltage, high-speed diodes are around 40 pF. When the FET switch570 is ON, the characteristics of the switch 570 in that state arerepresented as r_(on) resistance in parallel with the output capacitanceof the FET 570 as denoted as Co. The figure also shows packagingcapacitances Cpack1, Cpack2, which are discussed further below and aresimilar to the packaging capacitance Cpg discussed above.

When the FET switch 570 is in the ON state, the RF current through theswitch 570 must flow in both directions. Note that during the positiveswing of the current through the switch 570, current flows according topositive current path 558. At that time diodes 582, 584 are ON. Notethat the current in the FET 570 flows from drain 572 to source 573. Whenthe FET switch 570 is still in the ON state but the RF current flowsaccording to the negative current path 559, diodes 581 and 583 willconduct the RF current. Put more generically, it can be said that theswitching circuit comprises (a) a first switching device 581 that, onlywhen switched on, passes current from the second terminal 522 to thetransistor second terminal 572 (drain); (b) a second switching device582 that, only when switched on, passes current from the first terminal521 to the transistor second terminal 572 (drain); (c) a third switchingdevice 583 that, only when switched on, passes current from thetransistor third terminal 573 (source) to the first terminal 521; and afourth switching device 584 that, only when switched on, passes currentfrom the transistor third terminal 573 (source) to the second terminal522. While in the current embodiment the switching devices are diodes,in other embodiments they can other switching devices such astransistors (see, e.g., FIG. 32).

It is important to note that in both the negative current path 559 andthe positive current path 558, the current flows in the FET 570 fromdrain 572 to source 573. Thus, current flowing between the first andsecond terminals flows from the transistor second terminal 572 (drain)to the transistor third terminal 573 (source) both when the current isflowing in a positive direction and when the current is flowing in anegative direction. This is an important feature that prevents the needfor another FET transistor for the negative flow of RF current. In otherembodiments, positive and negative currents can instead flow from sourceto drain to prevent need for another FET transistor. In yet otherembodiments, a second FET transistor can be utilized.

FIG. 27 is a schematic representing the high-power RF switch 531 of FIG.25 in the ON state during the positive cycle, while FIG. 28 is aschematic representing the high-power RF switch of FIG. 25 in the ONstate during the negative cycle. These figures show the resistancer_(on) of the FET switch 570 as well as output capacitance Co of the FETswitch 570. The voltage drop across the resistance r_(on) will be verysmall if r_(on) has a value in few tens of milliohms, as is typical ofmost power transistors. Under these conditions, the output capacitanceCo is practically shorted out. The HV voltage drops from drain to groundor source to ground will be about the same. The nodes drain, and sourcewill have high voltages to ground but not from drain to source.

Condition When RF Switch is OFF

FIG. 29 is a schematic representing the high-power RF switch 531 of FIG.25 in the OFF state. As will be shown, the parasitic capacitance of theaccompanying diodes 581-584 play an important role. It is thereforeadvantageous for this application to find HV diodes with the lowestpossible capacitance and the lowest voltage drop when conducting RFcurrent from 10 to 15 A rms. There exist few such components now. Forinstance: CREE C3D10170H, I_(rf)=14 A rms, C_(diode)=40 pF at 1,000V.

Diode capacitances are shown in FIG. 29 as C_(D1) to C_(D4) belonging todiodes 581-584. In addition to diode capacitances, the figure shows thepackaging capacitance from source to ground Cpack2 and from drain toground Cpack1 on transistor 570. These parasitic capacitances are thecapacitances from the FET packaging to ground when the power FET ismounted on the heat sink with an electronically isolated and thermallyconducting pad. The combined capacitance of C_(D1) and Cpack1 can berepresented by C_(D1′), and the combined capacitance of C_(D4) andCpack2 can be represented by C_(D4′), whereC_(D1′)=C_(D1)//Cpack1C_(D4′)=C_(D4)//Cpack2

FIG. 30 is an equivalent circuit for the schematic of FIG. 29 with theaddition of a tuning circuit according to one embodiment. The firstequivalent circuit shows the combined capacitances for C_(D2) andC_(D3), as well as the combined capacitances for C_(D1′) and C_(D4′).The second equivalent circuit combines these capacitances to a singlecumulative parasitic capacitance Ceq. To determine Ceq, the followingequation can be used:

${Ceq} = \frac{\left( {C_{D\; 2} + C_{D\; 3}} \right)\left( {C_{D\; 1^{\prime}} + C_{D\; 4^{\prime}}} \right)}{C_{D\; 2} + C_{D\; 3} + C_{D\; 1^{\prime}} + C_{D\; 4^{\prime}}}$

As one can see, the total capacitance of all parasitic capacitances offour diodes reduces to about a value of one diode.

Tuning Out Parasitic Capaitance of Four Diodes

The equivalent circuit of FIG. 30 includes the parallel connection of atuning inductor. The value of the inductor is chosen such that it willresonate with the equivalent total capacitance Ceq of the diodes. Thisgives a high impedance at the frequency of operation, blocking RFcurrents from leaking through the switch in the OFF state.

Importantly, by this configuration, the parasitic capacitance Co of theswitching FET 570 is eliminated. In the OFF condition, the equationsshow that the voltage drop between the drain and source is the DCrectified voltage of the RF BUS 556, when the switching diodes are aboutmatched (to have the same parasitic capacitance). Also, the RF ACcurrent through the switching FET 570 in the OFF condition is zero.Since the voltage between terminals drain and source on switchingtransistor 570 is a direct current we do not expect any RF AC current toflow. The simulation of the circuit (discussed below) confirms theseresults. That is also a mechanism that vanishes the influence of theoutput capacitance of switching FET 570. In the OFF state there is no ACvariation on the terminals drain-to-source on switching transistor 570.Rectified DC voltage charges the Co capacitance of switching transistor570. If we put more of the same FET switches in series, that rectifiedDC voltage will distribute the voltage down evenly on each FET.

A practical calculation of the tuning inductor 579 in this case is asfollows. C_(D1)=C_(D2)=C_(D3)=C_(D4)=40 pF. Further, Cpack1=Cpack2=30 pFdistributed equally from drain to source (each node will see 15 pF).Putting these values into the equation above, Ceq=46.3 pF. Note that thepackaging capacitance of the switching transistor, for instance takenhere as 30 pF, contributes only 46.3 pF-30 pF, or 6.3 pF.

From the foregoing, it can be seen that one could use for a switchingtransistor a much less expensive FET since the FET's output parasiticcapacitance is cancelled and does not play any role in the circuit ifthe accompanying HV diodes are high speed diodes, and matched to havethe same parasitic capacitance. Thus, one can use a less expensive FETas a switching transistor if it also has low resistance r_(on) and highvoltage breakdown. The switching speed requirement is now reduced if itcan switch with the bandwidth of the power control loop. Thosefrequencies are, currently, less than 100 kHz or 10 us.

Tuning Inductor Value Calculation

The tuning out of the Ceq capacitance is accomplished by resonating withthe external inductance L between the terminals A and B. The inductanceis calculated using the following equation.

$L = \frac{1}{\omega^{2}{Ceq}}$

Where f=13.56 MHz and Ceq=46.3 pF, L=3 uH. This is a very reasonablevalue at that frequency and capacitance. Only one inductor will berequired to tune out the Ceq at high voltages on terminals A and B onthe RF switch.

FIG. 31 is a schematic providing a practical circuit layout for the forhigh-power RF switch of FIG. 25 according to one embodiment. It is notedthat the circuit has only five components and an inductor.

Second Embodiment (Using Transistors)

FIG. 32 is a second embodiment of the high-power RF switch 531-2utilizing transistors 581-2, 582-2, 583-2, 584-2 instead of diodesaccording to one embodiment. In this embodiment, the first transistor581-2 has a drain 581D coupled to the second terminal 522-2 and a source581S coupled to the transistor third terminal 573-2 (source); the secondtransistor 582-2 has a drain 582D coupled to the first terminal 521-2and a source coupled to a transistor third terminal 573-2 (source); thethird transistor 583-2 has a source 583S coupled to the first terminal521-2 and a drain 583D coupled to the transistor second terminal 572-2(drain); and the fourth transistor 584-2 has a source 584S coupled tothe second terminal 522-2 and a drain coupled to the transistor secondterminal 572-2.

In one embodiment, the transistors 581-2, 582-2, 583-2, 584-2 areinsulated-gate bipolar transistors (IGBTs), though the invention is notso limited. IGBTs have inherently larger break down voltages betweendrain and emitter contacts. A drawback for using such devices is thatthey are best at low frequency operation. The switch 531-2 would beespecially useful in a matching network at power levels up to 10 kW andfrequency below 500 kHz. The exemplified transistors 581-2, 582-2,583-2, 584-2 are synchronously driven every half cycle of theoperatizing frequency. The switching of ON/OFF of those transistorsshould still be very fast, so that in comparison to the diodes, in placeof transistors Q1 to Q4 the switching losses would be small. Thisarrangement as shown in FIG. 32 can have inherently much lower losseswhen compared to the diodes.

Like the first embodiment, this switch 531-2 includes a switchingcircuit 546-2 and a tuning circuit 547-2. The exemplified tuning circuitincludes a tuning inductor 579-2 and a tuning capacitor 578-2, thoughsimilar to switch 531 of the first embodiment, the tuning capacitor maybe omitted. The switch 531-2 shares many other features of switch 531.For example, positive and negative current both flow in the samedirection through the switching transistor 570-2. But by contrast, theswitching circuit 546-2 has three isolated gate drivers, specifically,two isolated gate drivers 575-2B, 575-2C for driving the transistors581-2, 582-2, 583-2, 584-2, and one isolated gate driver 575-2A fordriving the switching transistor 570-2.

Third Embodiment Higher (Voltage)

FIG. 33 is a third embodiment of the high-power RF switch 531-3 forincreased high voltage according to one embodiment. In this embodiment,each of the four switching devices 581-3, 582-3, 583-3, 584-3 comprisesa plurality of switching devices. In this embodiment, switching device581-3 comprises diodes 593-3, 594-3, switching device 582-3 comprisesdiodes 591-3, 592-3, switching device 583-3 comprises diodes 597-3,598-3, and switching device 584-3 comprises diodes 595-3, 596-3. Inother embodiments, transistors can be used instead of diodes. Further,in this embodiment the switching transistor 570-3 comprises a firsttransistor 570-3A and a second transistor 570-3B coupled in series.

This switch 531-3 is capable of switching 1,000V peak voltages andcurrents up to 50 A. The current capability and voltage capability willdepend upon the section of the FETs that are available now. Thesedevices, as of this writing, can withstand those voltages and currents.This switch will be useful in the design of a digital matching networkat a power level of P=10 kW and frequency of 13.56 MHz. The switch 531-3may also require a special gate driver design which is isolated andcapable of withstanding the isolation of HV transients at the gate up to10 kV. The gate driver can be a special and separate design in thiscase.

The diodes 591-3 to 598-3 of switch 531-3 are in series to increase thevoltage break down of the diode branch. The same is true for the FETtransistors 570-3A, 570-3B. Since they are in series, they can withstandhigher break down voltages. As shown in the simulation (below), when theRF switch 531-3 is in the OFF condition the RF AC current through theFETS 570-3A, 570-3B is zero. However, the voltage on the FETS 570-3A,570-3B from drain to source is a DC voltage rectified from the RF bussignal at the frequency of apportion and the power level. For instance,in the case of a P=10 kW matching network, based on 50 Ohms RF bus, thevoltage expected on the RF bus will be 1,000V peak. The same value of DCvoltage of 1,000V will appear on the terminals between the drain tosource on the FETs connected in series. Their DC voltage will distributeevenly and, so we expect that each FET will see only 500 V on itsterminals. That DC voltage build up between drain-to-source of theswitching FETs is also the reason why the tuning inductor 579-3 has adecupling capacitance, CL (some large values such as 1 nF/5 kV)associated with it. In the simulations circuits that capacitance has adesignation of C8.

As with the prior embodiments, switch 531-3 of FIG. 33 includes a firstterminal 521-3 and a second terminal 522-3, a tuning circuit 547-3comprising a tuning inductor 579-3 and an optional tuning capacitor578-3, and an isolated driver circuit 575-3 for driving the transistors570-3A, 570-3B.

Simulations

FIG. 34 is a simulation circuit for a high power-RF switch when in theOFF state according to one embodiment. The V1 input RF bus voltage is707.1 Vpeak, and Pin is 5 kW. Below is a table of the nodal analysis ofthe voltages and currents.

TABLE 3 Simulation Results Where Switch OFF left right delta min max ppmean rms acrms freq Screen 0    100e−6    100e−6 V(CCA) 0 1.935561.93556 −115.415 289.551 404.966 1.94186 9.3115 9.10677 12.5634e+6V(Rtest1) 0 776.546 776.546 0 808.16 808.16 792.066 792.175 13.125V(Rtest2) 0 −293.053 −293.053 −809.548 809.56 1.61911e+3 −1.93449498.959 498.956 13.5587e+6 V(D1) 0 −241.746 −241.746 −808.86 700e−3809.56 −395.066 467.281 249.548 13.5575e+6 V(D2) 0 −534.799 534.799−808.848 700e−3 809.548 −397 468.935 249.58 13.5595e+6 I(T1) 0 0 0 0 0 00 0 0 I(V1) 0 −7.36452e3  −7.36452e3  −9.75601 11.3973 21.15339.76269e−6 162.702e−3 162.702e−3 12.7862e+6 I(CCA) 0 −7.33541e−3−7.33541e−3 −9.75598 11.3973 21.1533 9.76195e−6 162.702e−3 162.702e−312.8062e+6 I(D1) 0 0 0 0 9.8919 9.8919 911.815e−6 79.8419e−3 79.8367e−34.21661e+6 I(D2) 0 0 0 0 9.81311 9.81311  1.1082e−3 88.0862e−388.0792e−3 6.14321e+6 I(D3) 0 0 0 0 9.81311 9.81311  1.1082e−388.0862e−3 88.0792e−3 6.14321e+6 I(D4) 0 0 0 0 9.8919 9.8919 911.815e−679.8419e−3 79.8367e−3 4.21661e+6 V(R1) 0 −291.117 −291.117 −712.739713.655 1.42639e+3 7.36896e−3 498.788 498.788 13.5593e+6

FIG. 35 is a simulation circuit for a high power-RF switch when in theON state according to one embodiment. Again, the V1 input RF bus voltageis 707.1 Vpeak, and Pin is 5 kW. Below is a table of the nodal analysisof the voltages and currents.

TABLE 4 Simulation Results Where Switch ON left right delta min max ppmean rms acrms freq V(CCA) −423.484 284.645 708.13 −686.076 686.0761.37215e+3 −614.893e−3 471.594 471.593 13.5593e+6 V(Rtest1) 2.3503361.6107 59.2603 −700.011 e−6 61.617 61.6177 21.7876 31.9252 23.334827.1186e+6 V(Rtest2) 3.75033 63.0107 59.2603 −63.017 63.017 126.034   313.43e−3 32.8964 32.8949 13.5593e+6 V(D1) −3.05033 −62.3107 −59.2603—62.317 700e−3 63.017 −11.0505 23.078 20.2603 13.5593e+6 V(D2) 700e−3700e−3 333.067e−18 −62.317 700e−3 63.017 −10.7371 22.7618 20.070313.5593e+6 I(T1) 23.5033 24.387 883.742e−3  −7.00011 e−3 24.8806 24.887618.7096 20.2587 7.76959 27.1186e+6 I(V1) 23.3276 24.2928 965.144e−3 −26.6369 26.6369 53.2738   226.809e−3 20.3227 20.3215 13.5593e+6 I(CCA)23.3277 24.2927 965.067e−3  −26.6369 26.6369 53.2738   226.809e−320.3227 20.3215 13.5593e+6 I(D1) 0 0 0 0 26.4599 26.4599 9.23378 14.276110.8879 13.5593e+6 I(D2) 23.4523 24.2826 830.337e−3  0 26.4599 26.45999.45966 14.4779 10.9602 13.5593e+6 I(D3) 23.4523 24.2826 830.337e−3  026.4599 26.4599 9.45966 14.4779 10.9602 13.5593e+6 I(D4) 0 0 0 0 26.459926.4599 9.23378 14.2761 10.8879 13.5593e+6 V(Ri) −419.734 347.656 767.39−687.498 687.498  1.375e+3 −301.463e−3 481.211 481.211 13.5593e+6

FIG. 36 is a waveform 588 of a signature of the RF current flowingthrough one of the discrete capacitors of the EVC according to oneembodiment. Specifically, the RF current is flowing through a discretecapacitor having a capacitance of 500 pF at 13.56 MHz that is beingswitched from OFF to ON. The calculated current in the matching networkis for a condition when the matching network operates at 5 kW (see tableabove). The waveform is that of discrete of capacitor current 20.3 Arms. The simulation levels for the above example were taken for P=5 kWat 13.56 MHz, and a 50Ω RF bus. We can achieve a discrete capacitorcapacitance of 500 pF, switching at P=5 kW match, in a shunt of thematching network with only one Power FET.

It is noted that switches discussed above can be used as part of avariety of matching networks, including the matching networks discussedin FIGS. 1A-10 discussed herein. The switches can also be used as partof a method for manufacturing a semiconductor as described herein.Further, the switches can form part of an EVC of a matching networkwhere the matching network forms part of a semiconductor processingtool, the semiconductor processing tool including a plasma chamber andan impedance matching network, such as any of the plasma chambers andimpedance matching networks described herein (e.g., matching network 100and load 120 of FIG. 1A). Further, while the switch is described asforming part of an EVC, the switch can also be used in otherapplications.

There are several advantages to the RF switches discussed above. Forexample, a matching network designed in pi configuration for a 5 kWapplication could use only one FET and four HV diodes as switches in thefirst shunt position. It will need only one parasitic capacitance tuningout inductor to tune out the parasitic capacitance. Further, theswitching FET can be any FET or any BJT. It can have relatively largeoutput capacitance (i.e., >200 pF). The output capacitance of aswitching FET in this switch can be cancelled out at the RF frequency.The FET parts are therefore also less expensive. Further, a totalfootprint of the layout is smaller. Further, the reflected HV transientsfrom the chamber get somewhat dissipated by the HV diodes. If the FETswitching transistor is in the OFF state, at that moment, the transientswill be seen on the diodes capacitances and those transient currentswill flow through the diode capacitance to ground. The FET switch issomewhat better protected by the switching diodes. Finally, higheroperation voltage is achieved by using the four diodes. Those fourdiodes at full load, conduct 41% smaller current than the switching FET.

RF Power Amplifier

As discussed above, there is need for a less expensive and moreefficient method for varying the output power of an RF generator. Belowwill be discussed a new circuit architecture that covers all RF poweramplifier classifications. The embodiments discussed will be class ABand class E RF amplifiers, but these are just examples and are notexhaustive. The new design does not need to vary the DC bus to vary thegain of RF power, nor does it need to adjust input power level, gatebias, or input phase. Rather, RF power can be controlled throughtransconductance variation. The RF power variation in this new circuitarchitecture is achieved by the variation of the drain current in activeswitching transistors (which may be several transistor in parallel forhigh power application) and hence variation of the transconductance ofthe RF power devices driving the amplifier chain. Note that anytransistor could be used here (BJTs, HEMTs, MOSFET, etc.), and MOSFETsare used simply as an example.

FIG. 37 is a block diagram of a topology for a universal RF amplifier2100 according to one embodiment. The amplifier includes three distinctRF power transistors blocks, namely, two RF power amplifier stages 2101,2102, and a transconductance control stage 2106, shown as a currentsource. This universal RF amplifier topology can be used for all classesof operation. If one changed the RF drive of the power amplifier blocks,one could invoke other classes of RF power amplifier design. At the sametime, if other classes of operation are required, one may change thedrain filters the power amplifiers 2101, 2102 through which the powergets delivered. For other classes of operation, the design may need agate biasing on RF stages such as a Class AB or C stage. In those cases,the system can bias the drain current to some small drain current in theorder of 0.1 A to 1 A.

The design allows for high frequency, arbitrary pulsing waveforms. Highfrequency pulsing with fast rise time can be achieved. The fast pulsing,with fast rise time, is inherent in this design since the RF current isnot interrupted. The RF current is either steered to the output networkof the amplifier or through the active switching devices. By starvingthe current in the power amplifiers 2101, 2102 and thus delivering lesscurrent to the output network, the amplifier 2100 controls the outputpower. By using this approach, the amplifier 2100 achieves fast risetimes in pulsing. The gates of power amplifiers 2101, 2102 are alwaysdriven by the constant RF drive.

As shown in FIG. 37, the power amplifiers 2101, 2102 are driven by an RFsource 2140. In the exemplified embodiment, the RF source 2410 iscoupled to an input 2112 of the amplifier 2100. The inputted RF signalis split by splitter 2110 coupled to gate drivers 2106, 2108, which inturn are coupled to the power amplifiers 2101, 2102. The poweramplifiers each output a signal to combiner 2104, which combines thesignal for the output 2114. A control circuit 2115 controls thetransconductance control stage 2106, thereby controlling the outputpower, as discussed below.

The RF source 2140 may be at any frequency, including from LF up tomicrowave. In the exemplified embodiment, the power amplifiers 2101,2102 are driven at a constant power, removing the need for a complexpreamplifier. In this embodiment all transistor gates are transformerisolated, because the DC supply is simply AC mains rectified. The RFsection is therefore floating. The gate transformers provide thegalvanic isolation. The gate drive on the transconductance controlcircuit 2106 is the control loop voltage that comes from the RF section.The envisioned control loop of the control circuit 2115 is analog but adigital control loop using an FPGA could be done as well. The controlboard in which the RF power sensor outputs meet will also have theserial port for the computer connection. Those standard techniques arenot discussed here. In other embodiments, the gates of the first andsecond transistors can be opto-isolated.

When designing such an amplifier at lower frequencies one could usesimply rectified 110 VAC, single-phase AC mains and achieve VDCrequired. There is no need for a sophisticated DC power supply, which isan important advantage of this design.

FIG. 38 is a schematic of an RF amplifier 2200 for a Class E amplifieraccording to one embodiment. Similar to FIG. 37, the amplifier 2200includes first transistor amplifier 2201, second transistor amplifier2202, and third transistor 2203 for controlling the first and secondtransistor amplifiers 2201, 2202. Further, the amplifier 2200 includesan RF source 2240, a gate driver 2206, output networks 2218, 2220, acombiner 2204 at the output, a transconductance control circuit 2222,and a control loop 2216 for RF power level control and pulsing.

The first and second transistors 2201, 2202 are RF power amplifierstages. The third transistor 2203 is a transconductance control stagefor both the first transistor 2201 and the second transistor 2202. Thethird transistor 2203 controls the current for both RF stages (first andsecond transistors 2201, 2202), as the current variation in the thirdtransistor 2203 directly impacts the transconductance of the firsttransistor 2201 and the second transistor 2202. The main DC bussupplying the amplifier stages is assumed constant (for example, about150 VDC). The values of DC voltage will depend upon the technology ofthe FETS and the frequency of operation. For instance, there are morehigh voltage FETS available at lower frequencies (≈10 MHz) than atfrequencies of 100 MHz or higher.

The third transistor 2203 (transconductance control stage) controls thedrain current to the first and second transistors 2201, 2202 (RF powerstages) if the control loop 2216 demands are set to deliver more or lesspower at the output. The current control stage (third transistor 2203)is driven by the control loop 2216 from the output of the RF poweramplifier. The process computer interfacing the load or process computersoftware could also drive the RF power loop. For instance, the systemmay dial into the power amplifier a certain power-preset point. That setpoint will bias the third transistor 2203 to set up the appropriate RFpower level for that process.

The drain current from the third transistor 2203 drives the draincurrents in the RF stages (first and second transistors 2201, 2202). Thedrain currents in the RF stages above share currents equally. The stagesare matched and then combined. In this embodiment of the architecture,there must be a dual pair of transistors in the RF stage, because onestage acts as a bypass of the source of the other at the source of theFET. Each source is bypassed by the low impedance of 1/g_(m) of theother stage (where g_(m) is the transconductance, and 1/g_(m) is inputresistance R_(in)). Since g_(m) is typically between 5 to 10 S for powerFETS, the 1/g_(m) term will give less than 1 Ohm of bypass resistance.That is sufficient to get large RF gain in each stage. Most of themodern power FETS available now are destined for high current switchingapplications. Therefore, we expect them to have relatively large valuesof transconductance (g_(m)). Some degeneration of the gain due to bypassing of 1/g_(m) of the next FET in parallel, connected at the source,is therefore tolerable. In fact, this technique is helpful to stabilizethe gain of the amplifier. By controlling the current in this way, thesystem does not interrupt the current flow in the choke inductors andfilters. The operation of these amplifiers should therefore be moreefficient under the abrupt output load impedance changes such as in theapplication in the semiconductor industry where such amplifiers aredriving the plasma loads during deposition or etching.

The following is an example of a 500 W design. During the simulationprocess, there is approximately 250 W per power FET when operated by+150 VDC fixed DC power bus. A typical one FET transistor load line isabout R_(FET)=3−j8Ω (at 60 MHz when the amplifier is driving the 50Ωinput of a matching network, for example). If the efficiency of the FETtransistor stage, in Class E mode, is assumed to be 85% or larger, thecalculated losses in one Power FET transistor are:

$\begin{matrix}{{P_{loss} = {P_{out}\frac{1 - \eta}{\eta}}}{P_{loss} = {250\frac{1 - 0.85}{0.85}}}{P_{loss} = {44\mspace{14mu}{W/{transistor}}}}} & (1)\end{matrix}$

An average RMS drain current per transistor is I_(D)=3 A rms, at anassumed frequency of operation of 60 MHz. A typical transconductance(g_(m)) of the FET amplifier during the transition when the transistorgoes from the ON to OFF or from the OFF to ON stage is shown in Equation2.

From the drain current (I_(D)) versus gate-to-source voltage (V_(GS)) ofa typical commercial FET we derived the transconductance (g_(m)) inTable 1 below. The values in the columns for g_(m) and R_(in) werecalculated from the data shown in the table. The FET transistorcharacteristics were taken from the data sheet for that device. Sometypical characteristics of a commercial FET are shown below. This devicewas used in the simulation process for this new circuit architecture.

TABLE 1 V_(GS) [V] I_(D) [A] g_(m) [S] R_(in) = (l/g_(m)) [M1 and M2] 3(V_(TH)) 0 — — 3.5 0.05 0.3 3.5 4 0.13 0.5 2 4.5 0.5 1 1 5 2 2 0.5 5.5 53.2 0.3 6 11 4.7 0.2 6.5 14 5.3 0.2

The transconductance calculation in Table 1 was calculated as shownbelow in one exampleg _(m)=2k(V _(gs) −V _(TH))  (2)Where:I _(D) =k(V _(gs) −V _(TH))²  (3)

This is just a regular MOSFET drain current in the linear region. Here,we assumed the threshold voltage (V_(TH)) is 3V for some FET and k=0.5[A/V], evaluated at V_(GS)=5V and I_(D)=2 A. The transconductance inTable 1 was calculated using Equation 2. From the transconductance(g_(m)), we also calculated the input resistance (R_(in)) looking intothe source of the RF stage either first transistor 2001 or secondtransistor 2002. In the control loop the V_(GS) of third transistor 2203is varied, thereby varying the I_(D) in first transistor 2001 and secondtransistor 2002.

In this example for the Class E amplifier, first and second transistors2201, 2202 its source is bypassed by input resistance R_(m). Asdiscussed, the active bypassing is helpful if the basic cell should havea large gain when switching to an ON or OFF state. Thus, in thisembodiment, it is required that at least two transistors (e.g., firstand second transistors 2201, 2202) be used and at least two transistorsmust be in the ON state at the same time to ensure 1/g_(m) (R_(in))bypassing.

In fact, it is this input resistance R_(in) that is varied by the thirdtransistor 2203, that in turn will be varied by the control loop 2216settings in current control stage. If the current in the thirdtransistor 2203 will be large (set by the controller), the g_(m) willalso be large, and hence the R_(in) will be small and the gain in thefirst and second transistors 2201, 2202 in the Class E stage will belarge. Varying the current in the third transistor 2206 varies the RFpower in the active RF stage of the first and second transistor 2201,2202.

FIG. 39 is a schematic of an RF amplifier 2300 for a Class AB amplifieraccording to one embodiment. It is clear that the fundamental structureof the basic cell of FIG. 37 is common between the Class E (FIG. 38) andClass AB (FIG. 39) examples. Transistors 2301, 2302, and 2303 aresimilar to transistors 2201, 2202, and 2203, respectively. Further, FIG.39 similarly includes an RF source 2340, a control circuit 2322, acontrol loop 2316, and a gate driver 2306. The amplifier 2300 furtherincludes a filter 2324 at the output.

The simplicity and flexibility of the topology, in addition with thereduction in necessary arts (as there is no need for complex DC suppliesor pre-amplifiers), is another advantage of this technique.

For higher power operation, or when using transistors with very largegain and high transconductance values, a quadrature gate drive can beused to improve performance. This ensures that current from the DCsupply to the load is never completely interrupted. The idea is thatthere will always be at least one power transistor in the ON state atany given time during operation.

FIG. 40 is a block diagram of a topology for a quadrature RF amplifier2400 for Class E operation according to one embodiment. In thisembodiment, the RF source 2440 provides an RF signal to a 90 degreesplitter 2410, which feeds signals to 180 degree splitters 2410A, 2410B,which feeds signals to gate drivers 2406, 2407, 2408, 2409, which feedsignals to amplifiers 2401, 2402, 2403, 2404, which output signals to180 degree combiners 2426, 2428, which feed signals to a 90 degreecombiner 2430, which provides a signal to the RF output. A controlcircuit 2415 is coupled to an optoisolated transconductance amplifier2406, whose drain is coupled to a source of each of the amplifiers 2401,2402, 2403, 2404 to control their output currents in a manner similar tothe method described above.

FIG. 41 provides a graph 2500 providing a detailed view of the timing ofthe gate drives according to one embodiment. As shown, during any givencycle, at least two transistors are fully ON. This avoids completelycutting off current from the DC bus. The quadrature gate drive alsohelps improve isolation between the RF sections, improving stability.This quadrature drive and combining can be achieved through both activeand passive networks. It gives a possibility of combing more power.

The pulsing of the RF power envelope may be achieved by modulating thecurrent in the transconductance control circuit. The pulsing rate willbe limited to less than 20/f₀, where f₀ is the RF operating frequency.For instance, if the operating frequency of the amplifiers is 60 MHz,the pulsing frequency can be extended up to about 3 MHz. Higher pulsingrates are possible, but aliasing becomes a problem. The third transistor2203 could be a bank of power FETS in parallel in order to deliversufficient current to the first and second transistors 2201, 2202 in oneexample.

In general, the circuit architecture discussed above is not frequencylimited. The RF power amplifier using this circuit architecture isviable from extremely low frequencies (less than 30 Hz) to microwavefrequencies (less than 3 GHz). Higher than 3 GHz is possible, but verydependent on device properties and expected efficiency would be low. Theproper selection of the active devices is critical. Active devices musthave large drain to source break down voltages, BV_(DS). The breakdownvoltages of transistors must be at least 3.5 times as large as V_(DC)selected for the Class E design for safe operation. Another reason forthat is that at some transient times, for example, the first or secondtransistors are in series with the third transistor and there exists apossibility to have a total DC bus voltage impressed upon each FETdevice.

The circuit architecture discussed herein may use 3 times as many powertransistors in comparison to other RF amplifier architectures. But todaypower transistors are reasonably priced and hence such circuits are veryeconomical. Since in this design there is no requirement for asophisticated DC power supply, the price offset of using more activedevices in RF stages is therefore appropriate. The DC bus for such RFarchitecture could simply be consisting of a rectified AC main with anappropriate filter capacitor.

The RF gate drives of the RF transistors can be standard, and thesedriver circuits are well known. The FET gates will be driven via theappropriate high frequency gate transformers since the galvanicisolation is required when the DC bus consists of a simple rectifiedmain.

The control loop will have wide bandwidth and should be fast. Thecontrol loop drives the transconductance control circuit to set thepower level and manage the amplifier to maintain steady output power.This control loop will be independent of the first and secondtransistors. The third transistor controls the current in the RF stagesof the first and second transistors. Therefore, the control loop willcontrol the currents in the first and second transistors. This controlloop will easily manage any ripple in the DC bus as well as voltagetransients on the DC bus caused by the main variations at 50/60 Hz. Inone embodiment, it is determined by simulation and will be better than1.25 μs response time, and have wide dynamic range.

It is noted that the RF power amplifier and method of amplifyingdiscussed above can be incorporated into any of the RF generators orother RF sources discussed above, including RF sources incorporated intoa system or method for manufacturing a semiconductor or other device. Inone method for manufacturing a semiconductor, a substrate is placed inthe plasma chamber, the plasma chamber configured to deposit a materiallayer onto the substrate or etch a material layer from the substrate.The plasma in the plasma chamber is energized by coupling RF power froman RF source into the plasma chamber to perform a deposition or etching.While energizing the plasma, the matching network between the RF sourceand plasma chamber can carry out an impedance match. The RF source cancomprise an RF power amplifier as discussed herein. The semiconductormanufacturing system can comprise a plasma chamber and an RF source,such as any of the plasma chambers and RF sources described herein(e.g., load 120 and RF source 110 of FIG. 1A).

As discussed above, the above approach is a less expensive system designfor RF power amplifiers. There is no need for a complex DC power supplywith varying DC levels or complex preamplifiers. Therefore, much lessweight, space, and thermal control will be needed. Arbitrary pulsing atfrequencies and waveforms can be achieved. The RF generators will becomemore reliable through component reduction and system simplification.Pulsing would be improved in all generators and at all frequencies. Thesame basic cell RF architecture could be used in all products and forall frequencies. There is no need to have different circuit architecturefor different frequencies. Large savings could be achieved by buyingcommon parts. Further, service technicians can more easily make repairs.

For the circuits discussed above, note that not all components areshown. Rather, the drawings show those components helpful in conveyingan understanding of the circuit and its operation. A person of ordinaryskill in the art will readily understand the role other standardcomponents can play in the circuit. Further, it is noted that the aboveswitching circuits can be used in methods for providing switching,including methods to provide switching to capacitors or inductors in amatching network, including matching networks in semiconductorfabrication. It is further understood that where devices are describedas being operably coupled or coupled, such coupling may be direct orindirect.

While the inventions have been described with respect to specificexamples including presently preferred modes of carrying out theinvention, those skilled in the art will appreciate that there arenumerous variations and permutations of the above described systems andtechniques. It is to be understood that other embodiments may beutilized and structural and functional modifications may be made withoutdeparting from the scope of the present inventions. Thus, the spirit andscope of the inventions should be construed broadly as set forth in theappended claims.

What is claimed is:
 1. A radio frequency (RF) power amplifiercomprising: a first transistor and a second transistor in parallel,wherein a gate of the first transistor and a gate of the secondtransistor are configured to be driven by an RF source; a thirdtransistor comprising a drain operably coupled to both a source of thefirst transistor and a source of the second transistor; atransconductance control circuit operably coupled to a gate of the thirdtransistor and configured to alter a gate-to-source voltage of the thirdtransistor, thereby altering a drain current of each of the firsttransistor and the second transistor, thereby altering an output powerof the RF power amplifier; and a control loop configured to drive thetransconductance control circuit.
 2. The amplifier of claim 1 whereinthe first, second, and third transistors are MOSFETs.
 3. The amplifierof claim 1 wherein the altering of the gate-to-source voltage of thethird transistor alters a drain current of the third transistor, therebyaltering the drain current of the first and second transistors.
 4. Theamplifier of claim 1 wherein the RF source drives the first and secondtransistors at a constant power.
 5. The amplifier of claim 1 wherein thegates of the first and second transistors are transformer isolated oropto-isolated.
 6. A method of amplifying RF power, the methodcomprising: coupling a first transistor and a second transistor inparallel; driving, by an RF source, a gate of the first transistor and agate of the second transistor; coupling a drain of a third transistor toboth a source of the first transistor and a source of the secondtransistor; coupling a transconductance control circuit to a gate of thethird transistor; driving the transconductance control circuit by acontrol loop; and altering, by the transconductance control circuit, agate-to-source voltage of the third transistor, thereby altering a draincurrent of each of the first transistor and the second transistor,thereby altering an output power of the RF power amplifier.
 7. Themethod of claim 6 wherein the first, second, and third transistors areMOSFETs.
 8. The method of claim 6 wherein the altering of thegate-to-source voltage of the third transistor alters a drain current ofthe third transistor, thereby altering the drain current of the firstand second transistors.
 9. The method of claim 6 wherein the RF sourcedrives the first and second transistors at a constant power.
 10. Themethod of claim 6 wherein the gates of the first and second transistorsare transformer isolated or opto-isolated.
 11. A method of manufacturinga semiconductor, the method comprising: placing a substrate in a plasmachamber configured to deposit a material layer onto the substrate oretch a material layer from the substrate; energizing plasma within theplasma chamber by coupling RF power from an RF source into the plasmachamber to perform a deposition or etching, the RF source comprising anRF power amplifier comprising: a first transistor and a secondtransistor in parallel, wherein a gate of the first transistor and agate of the second transistor are configured to be driven by an RFsource; a third transistor comprising a drain coupled to both a sourceof the first transistor and a source of the second transistor; and atransconductance control circuit coupled to a gate of the thirdtransistor; and a control loop configured to drive the transconductancecontrol circuit; and altering, by the transconductance control circuit,a gate-to-source voltage of the third transistor, thereby altering adrain current of each of the first transistor and the second transistor,thereby altering an output power of the RF power amplifier.
 12. Themethod of claim 11 wherein the altering of the gate-to-source voltage ofthe third transistor alters a drain current of the third transistor,thereby altering the drain current of the first and second transistors.13. The method of claim 11 wherein the RF source drives the first andsecond transistors at a constant power.
 14. The method of claim 11wherein the gates of the first and second transistors are transformerisolated.